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IC ^°^^ 
 
 Bureau of Mines Information Circular/1985 
 
 A Low-Cost FSK Modem Network 
 for Polled Communication Systems 
 
 By Richard A. Watson 
 
 UNITED STATES DEPARTMENT OF THE INTERIOR 
 
 1751 
 
 'Wines 75TH ax^"^ 
 
i 
 
Information Circular 9021 
 
 A Low-Cost FSK Modem Network 
 for Polled Communication Systems 
 
 By Richard A. Watson 
 
 UNITED STATES DEPARTMENT OF THE INTERIOR 
 Donald Paul Model, Secretary 
 
 BUREAU OF MINES 
 Robert C. Horton, Director 
 
Libtaty of Congress Cataloging in Publication Data: 
 
 no. ^0^1 
 
 Watson, Richard A., 1948- 
 
 A low-cost FSK modem network for polled communication systems. 
 
 (Information circular ; 9021) 
 
 Includes bibliographical references, 
 
 Supt. of Docs, no.: I 28.27:9021. 
 
 1. Mine communication systems. 2. Modems. I. Title. II. Title: 
 Low-cost F.S.K. modem network for polled communication systems. III. 
 Series: Information circular (United States. Bureau of Mines) ; 9021. 
 
 -T!<r29-5rU4— [TN3441 622.s [622', 81 
 
 84-600286 
 
 V^^" 
 
 v> 
 
 < 
 
CONTENTS 
 
 Page 
 
 Abstract I 
 
 Introduction 2 
 
 Purpose and requirements 2 
 
 General description 3 
 
 Theory of operation 4 
 
 Basic overview 5 
 
 LSM operation 6 
 
 RCCM operation 9 
 
 Other applications 11 
 
 Conclusion. 12 
 
 Appendix A. — Frequency selection and filter design 13 
 
 Appendix B. — Modulation design and calibration 18 
 
 ILLUSTRATIONS 
 
 1 . Modem frequency spectrum 4 
 
 2 . Network diagram 4 
 
 3 . LSM block diagram • 3 
 
 4 . RCCM block diagram 6 
 
 5 . Waveform timing diagram 6 
 
 6. LSM schematic 7 
 
 7. Line splitter schematic. 9 
 
 8. RCCM schematic 10 
 
 A-1. Chebyshev bandpass filter 14 
 
 A-2. Filter response 16 
 
 A-3. RCCM filtered response 17 
 
 B-1. Demodular circuit 18 
 
 B-2. Modulator circuit 19 
 
 B-3. RCCM calibrator schematic 21 
 
 B-4. LSM printed circuit layout 22 
 
 B-5. Line splitter printed circuit layout 23 
 
 B-6. RCCM printed circuit layout 23 
 

 UNIT OF MEASURE 
 
 ABBREVIATIONS 
 
 USED 
 
 IN THIS 
 
 REPORT 
 
 Bd 
 
 baud (bits per second) 
 
 
 Mfi 
 
 megohm 
 
 dB 
 
 decibel 
 
 
 
 
 ms 
 
 millisecond 
 
 dB/mi 
 
 decibel per mile 
 
 
 
 yF 
 
 microfarad 
 
 ft 
 
 foot 
 
 
 
 
 ys 
 
 microsecond 
 
 h 
 
 hour 
 
 
 
 
 a 
 
 ohm 
 
 Hz 
 
 hertz 
 
 
 
 
 pF 
 
 picofarad 
 
 kHz 
 
 kilohertz 
 
 
 
 
 V 
 
 volt 
 
 kn 
 
 kilohm 
 
 
 
 
 w 
 
 watt 
 
 MHz 
 
 megahertz 
 
 
 
 
 
 
A LOW-COST FSK MODEM NETWORK FOR POLLED COMMUNICATION SYSTEMS 
 
 By Richard A. Watson' 
 
 ABSTRACT 
 
 A frequency-shift keying (FSK) modulator-demodulator (modem) network 
 has been devised for the Bureau of Mines mine-monitoring systems. This 
 network permits an unlimited number of remote stations to communicate 
 with one central station over the same pair of wires. The network oper- 
 ates at a speed of 4,800 Bd at distances up to 5 miles. 
 
 This report describes the development and operation of the modems. 
 The theory of operation, schematic diagrams, and printed circuit board 
 layouts are provided. Although the modems were built for a specific ap- 
 plication, a more general use is discussed whereby the network would op- 
 erate with a typical serial channel format. 
 
 'Electrical engineer, Pittsburgh Research Center, Bureau of Mines, Pittsburgh, PA. 
 
INTRODUCTION 
 
 The mining industry is finding that the 
 use of monitoring systems can increase 
 safety and production. Remote monitoring 
 gives management real-time information 
 about the underground environment and 
 equipment status. It also provides the 
 ability to remotely control equipment 
 such as belts and pumps. With this in- 
 formation at hand, hazardous conditions 
 and equipment failures are quickly recog- 
 nized and appropriate actions are more 
 immediate. 
 
 As the benefits from these monitoring 
 systems become documented, their growth 
 can be expected to increase; with this, 
 the distance and number of monitoring and 
 control functions will increase. Two im- 
 portant criteria must be considered: 
 system speed and data security. 
 
 The Bureau has researched monitoring 
 systems for the purpose of establishing 
 performance guidelines. Out of this re- 
 search, a special FSK modem has been de- 
 vised for use in the Bureau's monitoring 
 systems. This modem permits operation 
 over standard telephone wiring at dis- 
 tances up to 5 miles, with no limit on 
 the number of remote outstations. It 
 maintains the high speed of the baseband 
 link used for shorter distances and pro- 
 vides for lower error rates. 
 
 Although this modem was developed for a 
 special application that may be limited, 
 another application has been considered. 
 It entails modifying the modem to operate 
 with a typical asynchronous 11-bit serial 
 channel. This modification is being 
 tested and is discussed in some detail in 
 the section "Other Applications." 
 
 PURPOSE AND REQUIREMENTS 
 
 The existing communication link used in 
 the Bureau's monitoring projects consists 
 of baseband signals transmitted over a 
 special heavy conductor cable. Two of 
 these cables are used in the intrinsi- 
 cally safe mine-monitoring system (ISMMS) 
 installed at the Lucerne No. 8 Mine, 
 Clarksburg, PA. 2 Each cable is used to 
 communicate with and power environmental 
 transducers. 
 
 When expansion of the belt-monitoring 
 system was planned, another method of 
 communication to the underground was de- 
 sired. Modems were considered, because 
 they would employ available pairs of wir- 
 ing in the underground telephone network. 
 This would eliminate the costs and ef- 
 forts of installing more cable. The spe- 
 cial cable was not required since belt 
 monitoring was not included in intrinsic 
 safety operation. 
 
 ^Watson, R. A. An Intrinsically Safe 
 Environmental Monitoring System for Coal 
 Mines. Paper in Proceedings of the Sixth 
 WVU Conference on Coal Mine Electrotech- 
 nology (Morgantown, WV, July 28-30, 
 1982). WV Univ., Morgantown, WV, 1982, 
 pp. 345-360. 
 
 There are about 15 belt drive control- 
 lers in the Lucerae No. 8 Mine. With the 
 addition of several pump-monitoring sta- 
 tions, it was obvious that the number of 
 remote modems should not be limited. 
 Each remote modem must connect with the 
 local modem at the central station on the 
 surface. The modems should be transpar- 
 ent to the monitoring system, receiving 
 and transmitting compatible signals at 
 each end. 
 
 The picture that was rapidly developing 
 was of a system similiar to that of local 
 networks. With local networks, many us- 
 ers have the ability to share the re- 
 sources of a centralized computer via a 
 common communication link. Some networks 
 use carrier sense multiple access with 
 collision detection (CSMA/CD) and rely 
 upon remotes' transmitting on a random 
 basis, with provisions for detecting col- 
 lisions. Other networks use a token 
 passing scheme where information is sent 
 around a loop and each station takes or 
 injects what it needs. These, as well as 
 other types of networks, are complicated, 
 costly, and not directly compatible with 
 the polling scheme used in the Bureau's 
 monitoring systems. 
 
In order to have an unlimited number of 
 remote modems connected onto the same 
 line, each remote must be seen as a high 
 impedance. The receiving section should 
 be high-impedance-buffered, and the 
 transmitting section should be discon- 
 nected when not required to transmit. 
 The line is then terminated, at the re- 
 spective ends, with a resistance equal to 
 the characteristic impedance of the 
 line. 
 
 Both local and remote modems should be 
 guarded against electrical noise, spikes, 
 and surges on the line. The degree of 
 protection should be high, as mine elec- 
 trical disturbances are known to be con- 
 siderable. Additionally, the local modem 
 connection to the monitoring system 
 should be optically isolated from lines 
 that enter the underground. 
 
 GENERAL DESCRIPTION 
 
 An FSK modulator converts binary digi- 
 tal values, representing logical "ones" 
 (1) and "zeros" (0), into two frequen- 
 cies. They are known as the mark and 
 space frequencies, respectively. The de- 
 modulator reconverts these frequencies 
 into digital values at the receiving end. 
 Exar Integrated Systems, Inc.,3 manufac- 
 tures integrated circuits (IC's) that do 
 the conversions with a small number of 
 external components. Two of these IC's 
 were selected because of their low cost, 
 ease of use, and well-documented applica- 
 tion notes. They are the XR2206 modula- 
 tor and XR2211 demodulator. The pair can 
 accommodate signals over a 60-dB dynamic 
 range of frequencies as high as 300 kHz. 
 
 The modulator and demodulator IC's are 
 used in both the local and remote modems. 
 In the local modem, the carrier is always 
 on. A crystal-controlled clock synchro- 
 nizes timing with the transmitted message 
 and "opens" the receiver when the receiv- 
 ed message is due. This modem will be 
 referred to as the local synchronous 
 modem (LSM) . In the remote modem, the 
 carrier is turned on only when activated 
 by the station responding through the 
 modem. This modem will be referred to as 
 the remote carrier-controlled modem 
 (RCCM). 
 
 Although the mark and space frequencies 
 of the modem IC's cover a wide range, the 
 choice is not completely arbitrary. 
 
 ^Reference to specific manufacturer or 
 trade names is for identification only 
 and does not imply endorsement by the 
 Bureau of Mines . 
 
 Selection of these frequencies is based 
 upon the channel bandwidth, line attenua- 
 tion, and the system baud rate. Addi- 
 tionally, if a full duplex channel (simu- 
 ltaneous transmission in both directions 
 at the same time) is selected, the mark 
 and space frequencies of the two groups 
 must be separated by enough distance to 
 prevent interference. This separation is 
 determined based on the filtering re- 
 quirements of each group. 
 
 Figure 1 is an illustration of the two 
 groups of frequencies. The mark and space 
 frequencies transmitted from the LSM are 
 fl and f2, respectively. Those received 
 from the RCCM are fj, and f/^. The selec- 
 tion of these frequencies and design sol- 
 utions are covered in the appendixes; the 
 frequencies are given here as 12.6, 16.2, 
 23.4, and 27.0 kHz, respectively. The 
 selection avoided too high a frequency 
 where line attenuation is greater and 
 line balancing and matching become 
 important. 
 
 Separate pairs of wires are used for 
 the transmit and the receive frequencies. 
 A line hybrid could combine the two 
 groups onto the same line but would be 
 incompatible with the switching technique 
 of the transmitter. A hybrid is used to 
 balance a line and prevent the transmit- 
 ted output from feeding back into its re- 
 ceiver input. The balance is based on a 
 fixed line impedance as seen by both the 
 output and input and is used where modems 
 are connected one to one. When many re- 
 mote modems are connected to the same 
 line and are being switched on and off, 
 the line impedance is not constant. 
 
f 
 
 '2 '3 
 
 FREQUENCY SPECTRUM 
 
 FIGURE 1. - Modem frequency spectrum. (H(jw) = frequency response of a network function; H = 
 frequency response in the bandpass.) 
 
 With separate transmit and receive 
 lines, it is not neccessary that the two 
 groups of frequencies be separated, so 
 long as line and circuit crosstalk are 
 low. However, given the rugged environ- 
 ment of the mining industry, the line 
 conditions cannot be guaranteed. Bad 
 
 splices and line leakage could be expect- 
 ed with time. Because of these consider- 
 ations, the groups were separated and 
 filters were included at each receiver to 
 guarantee at least 24-dB attenuation to 
 out-of-band signals. 
 
 THEORY OF OPERATION 
 
 Operation of the modems requires one 
 LSM modem connected to at least one RCCM 
 modem, a 4,800-Bd serial computer channel 
 (monitoring system) , and a remote trans- 
 ponder. ^ The monitoring system communi- 
 cates two 11-bit words (back to back) to 
 the transponder through the LSM and RCCM, 
 as shown in figure 2. The transponder 
 then enables the carrier of the RCCM and 
 communicates its reply of two 11-bit 
 words to the monitoring system through 
 the RCCM and LSM. There are no limits to 
 the number of RCCM's connected to the LSM 
 as each are seen as a high impedance. 
 The number of transponders connected to 
 each RCCM can also be quite large and is 
 
 ■^The transponder used in the Bureau's 
 monitoring system is a special device. 
 Upon recognition of a given address and 
 command, it issues a synchronizing pulse 
 followed, 1 1 bit times later, with two 
 data words. It is essential to the oper- 
 ation of the LSM and RCCM in the original 
 development. 
 
 limited only by the supply current, line 
 length, and definition capabilities of 
 the monitoring system. 
 
 LSM 
 
 Terminators 
 
 4-wlre cable 
 2 transmit 
 2 receive 
 
 RCCM 
 
 ^, 
 
 © ® 
 
 RCCM 
 
 ^ 
 
 <::> 
 
 I 
 
 4-wire cable 
 
 +V 
 
 -V 
 Transmit 
 Receive 
 
 MS 
 
 LSM 
 RCCM 
 
 KEY 
 Monitoring system or any 
 serial computer channel 
 Local synchronous modem 
 Remote carrier-controlled 
 modem 
 Transponder 
 
 FIGURE 2. - Network diagram. 
 
BASIC OVERVIEW 
 
 Figure 3 is a block diagram of the LSM. 
 Communication between the monitoring sys- 
 tem and the LSM is by way of RS232 speci- 
 fied connections. Optical couplers pro- 
 vide electrical isolation of the signals. 
 The transmitted signal is passed to the 
 modulator and the synchronous circuit. 
 Mark and space frequencies from the modu- 
 lator (f ] and £2) are coupled to the line 
 through a transformer and line protection 
 circuit. Precise timing of the synchro- 
 nous circuit opens the channel gate when 
 the transponder's reply is due. 
 
 The responding mark and space fre- 
 quencies (f3 and f^) from the RCCM are 
 coupled to the demodulator through the 
 transformer and line protection circuit 
 and a bandpass filter. A reference 
 oscillator (at frequency f ^) keeps the 
 demodulator chatter down should a 
 transponder fail to respond in the re- 
 quired timeframe. The demodulated 
 signals signify reception to the 
 synchronous circuit and are then passed 
 
 through the gate to the monitoring 
 system. 
 
 Figure 4 is a block diagram of the 
 RCCM. The mark and space frequencies 
 from the LSM (f , and f2) are coupled to 
 the demodulator through a transformer and 
 line protection circuit, a high-impedance 
 buffer, and a bandpass filter. Demod- 
 ulated signals are then passed directly 
 to the transponders. The original RCCM, 
 as given in the section "RCCM Operation," 
 relies on the transponders to issue a 
 synchronizing pulse that triggers the 
 carrier and to delay the data to be 
 transmitted. The updated version, 
 discussed in the section "Other Applic- 
 ations," provides the trigger and delay 
 directly on the RCCM. Although this 
 increases the parts count and complexity 
 of the RCCM, it provides for a more 
 general use. 
 
 Modulated data from the RCCM (fj and 
 f ^) are passed to the line through a 
 solid-state switch and transformer and 
 line protection circuit. The switch pro- 
 vides greater than 50 dB of isolation in 
 
 Mine 
 
 monitoring 
 
 system 
 
 CPU 
 
 TXD 
 
 COMM 
 
 RXD 
 
 Optical 
 coupler 
 
 * -> 
 
 Synchronous 
 circuit 
 
 Optical 
 coupler 
 
 Modulator 
 
 Reference 
 oscillator 
 
 Gate 
 
 Dennodulator 
 
 Trans- 
 former 
 and 
 line 
 protection 
 
 Bandpass 
 filter 
 
 Line 
 out 
 
 FIGURE 3. - LSM block diagram. 
 
Line in- 
 
 Line out- 
 
 Transformer 
 
 and line 
 
 protection 
 
 
 High- 
 impedance 
 buffer 
 
 ^- 
 
 Bandpass 
 filter 
 
 
 Demodulator 
 
 — ►D 
 
 ^ 
 
 
 
 
 
 
 
 
 
 
 
 
 Carrier 
 control 
 
 
 
 
 
 
 
 * 
 
 
 
 
 
 ^ 
 
 
 .^ 
 
 Switch 
 
 ^ 
 
 Modulator 
 
 ^ 
 
 
 ^ Delay ^ 
 
 
 
 
 * 
 
 
 
 
 
 Data in 
 
 FIGURE 4. - RCCM block diagram. 
 
 the OFF state and less than 0.3-dB loss 
 in the ON state. Thus, the line is not 
 loaded by the output impedance of the 
 modulator when not in the transmit mode. 
 
 LSM OPERATION 
 
 This discussion deals with the original 
 LSM. Refer to the timing diagram of fig- 
 ure 5 and the schematic diagram of figure 
 6. The updated version under test is 
 discussed in the section "Other Applica- 
 tions." The appendixes can be consulted 
 for specifics on the operation of the 
 modulator, demodulator, and filter. 
 
 The transmitted signal enters the LSM 
 from the monitoring system at connector 
 Jl pin 3 or from a portable tester at J2 
 pin 2. Waveform 1 of figure 5 shows the 
 transmitted signal as seen at test point 
 TP2. This signal enters the modulator, 
 IC2, at pin 9. Potentiometers R55 and 
 R57 adjust the frequencies f, and f2. 
 R61 sets the output amplitude to dB as 
 measured at TP3. Diodes D3 and D4 and 
 resistor R9 provide line protection for 
 surges below the threshold of the gas 
 discharge tube on the secondary side of 
 transformer T2. 
 
 The transmitted signal also enters the 
 synchronous circuit at IC7 pin 6. Pin 5 
 is high at bit time (beginning of 
 
 I I I I I I I I l | I I I I I I I I l| I I I I I I I I l | I I I I I I I I l | I I I I I I I M | I M I I I I I I 
 
 Transmit 1 1 Transmit 
 
 
 Receive 
 I 
 
 Receive 
 2 
 
 I 1 1 1 1 1 1 I 
 
 ill,,, 
 
 20 30 
 
 TIME DIAGRAM, 
 
 40 
 bit time 
 
 50 
 
 60 
 
 FIGURE 5. - Waveform timing diagram. 
 
 I 
 
u 
 
 LU 
 
 Z) 
 
 o 
 
transmitted message) and can be measured 
 at TP7. Waveform 4 is the signal at TP7 
 and is known as the channel-open pulse. 
 With coincidence of the transmitted sig- 
 nal, a flip-flop of IC5 is set and pin 2 
 goes low as shown in waveform 2. 
 
 An oscillator circuit, running at 
 2.4576 MHz, is used as a clock input to 
 dividers IC3, IC4, and IC,3. With a low 
 on pin 11, IC3 begins to count clock pul- 
 ses until bit time 16 occurs. At this 
 time, a high at pin 3, waveform 3, sets 
 two flip-flops of ICg at pins 6 and 8. 
 ICg pin 2 goes low and thus opens the 
 channel for reception and blocks the 
 transmitted message from setting IC5 with 
 data pulses. ICg pin 12 also goes low, 
 as shown in waveform 5, and begins the 
 count of the time-out timer, IC13. 
 Shortly after bit time 16, IC7 pin 3 goes 
 high and resets ICg at pin 4. Meanwhile, 
 IC , 3 continues counting for 16 more bit 
 times and then produces a time-out pulse 
 at pin 3. This pulse is shown in wave- 
 form 6 and is the reset to ICg pin 10. 
 
 At bit time 16, the channel is open 
 (waveform 4) and the RCCM should be re- 
 ceiving the second word of the message 
 shown in waveform 1. At the completion 
 of this message, the reply from the RCCM 
 is expected. 
 
 The reply from the RCCM can be monitor- 
 ed at TP4. Dl, D2, and RIO provide low- 
 level protection for the received lines. 
 After passing through the buffer and 
 bandpass filter of IC,2j the mark and 
 space frequencies are delivered to the 
 demodulator, IC^. R27 adjusts the center 
 frequency between f3 and f^, D5 indi- 
 cates a carrier level sufficient for 
 reception. 
 
 ICj4 is another modulator set to oper- 
 ate at the receive space frequency, f/^. 
 R38 adjusts the frequency, and R39 is 
 adjusted to provide about a -50-dB level 
 as measured at TP9. This modulator 
 serves as a reference oscillator to the 
 phase-locked loop of the demodulator. It 
 prevents chatter from appearing at the 
 demodulated data output, pin 7, when no 
 response is received and the channel is 
 open. Normally, pin 6, tied to pin 7, 
 holds the data output off when no carrier 
 is present at the input, pin 2. However, 
 
 the tight bandwidth of the filter atten- 
 uates out-of-band noise and increases the 
 probability of momentary phase lock on 
 in-band noise and, thus, data chatter. 
 
 The demodulated data output is measured 
 at TP6 and shown as waveform 7. The 
 pulse at bit time 10.5 is intermittent 
 and occurs about half the time. It is a 
 result of the carrier's being just turned 
 on at the remote RCCM. Although the RCCM 
 turns on at the same frequency as the re- 
 ference oscillator (f4), phase differ- 
 ences cause the momentary pulse. This 
 pulse is the primary reason for the 
 synchronized channel-open pulse. Typical 
 modems, connected one to one, usually 
 have this pulse only during connect time. 
 It is commonly known, and software proto- 
 cols account for its presence. As there 
 was no control over the software for this 
 network, the solution was provided in 
 hardware. 
 
 The demodulated data at TP6 are coupled 
 to the monitoring system through Jl pin 
 2, or to the portable tester at J2 pin 1, 
 and are also fed to the synchronous cir- 
 cuit. It is logically OR'd with the 
 time-out pulse and logically AND'd with 
 the Inverse channel-open pulse at ICg pin 
 1. The resulting pulse sets IC5 flip- 
 flop at pin 8. A low at IC5 pin 12 en- 
 ables IC4 to count for 24 bit times. At 
 the end of that time, IC7 pin 10 goes 
 high and resets the counter and channel- 
 open pulse. Waveform 8 is actually the 
 Inverse of the 24-bit time at pin 12 of 
 IC5, and shows the OR'lng of waveforms 6 
 and 7 to set the time for waveform 4, the 
 channel-open pulse, to end. The synchro- 
 nous circuit is now reset and awaits an- 
 other transmission. 
 
 For implementation at the Lucerne No. 8 
 Mine, the transmitted signals needed to 
 be split into two directions, one for the 
 south mains line and one for the north. 
 To accomplish this, the circuit of figure 
 7 was connected between the LSM and the 
 RCCM's. The gas discharge tubes were 
 left off the secondaries of transformers 
 Tl and T2 of the LSM. Instead, the 
 transformers were connected to Tl and T2 
 of the line splitter, and here the gas 
 tubes were placed on the secondaries. 
 The transmitted signal is equally split, 
 
TBI-I > 
 
 TXD 
 
 TBI-2> 
 
 Shield TBi-3> 
 
 TBl-5> 
 
 Tl 3 
 
 G1 
 
 I 
 
 G2 
 
 $ 
 
 G3 
 
 I 
 
 G4 
 
 5 
 
 Vv^A 
 
 R2 
 -W\A- 
 
 R1 
 
 R4 
 
 R3 
 
 -v\AA 
 
 R6, , 
 -^AAA- 
 
 R5 
 
 R8 
 
 — nAAA- 
 
 R7 
 
 < TB2-I 
 
 < TB2-2 
 
 < TB2-9 
 
 North TXD 
 
 South TXD 
 
 < TB2-8 
 
 < TB2-5 Shield 
 
 < TB2-6 
 
 < TB2-7 
 
 < TB2-4 
 
 South RXD 
 
 < TB2-3 
 
 North RXD 
 
 Material list 
 
 Resistors R1-R8: 10ii,2W 
 Gas discharge tubes G1-G4: Tl I 339 
 
 Transformers T1,T2: TRIAD SP-69 
 Terminal board TB1: RDI3PCV-05 
 Terminal board TB2: RDI3PCV-09 
 
 FIGURE 7. - Line splitter schematic. 
 
 and the 600-fi output impedance of the LSM 
 modulator is transformed to 150 fi. This 
 impedance more accurately reflects that 
 seen in underground wiring. Since all 
 RCCM's reflect a high impedance, a 150-fi 
 resistor is connected at the end of the 
 transmit lines in both the north and 
 south mains. 
 
 RCCM OPERATION 
 
 The following discussion deals with the 
 operation of the RCCM. Refer to the 
 schematic of figure 8 and waveforms of 
 figure 5. 
 
 The mark and space frequencies from the 
 LSM are received through transformer Tl. 
 Surge protection is provided by the same 
 methods as for the LSM. The received 
 message is delivered to the demodulator, 
 IC,,. through the buffer and bandpass fil- 
 ter of IC5. R24 adjusts the center fre- 
 quency between f j and f2. D6 indicates a 
 
 received carrier of sufficient level for 
 reception. The demodulated data are 
 passed to the transponder through Jl pin 
 2 and can be measured at TP3 as that of 
 waveform 1. 
 
 The reply from the transponder can be 
 measured at TP4 and is the inverse of 
 waveform 7. The pulse at bit time 10.5 
 triggers IC4 at pin 2. This trigger 
 causes pin 3 to go high and results in 
 DIO going off and IC3 going on. IC3 is a 
 solid-state field effect transistor (FET) 
 switch with four stages in parallel. The 
 total ON resistance is about 10 Q.. At 
 this moment, the carrier is on and t^ is 
 being transmitted back, to the LSM through 
 T2. IC3 will remain on until IC4 times 
 out. This time is controlled by R26 and 
 C16 and is approximately 9 ms. This is 
 enough time, including component toler- 
 ances, to leave the carrier on for com- 
 plete transmission of the transponder's 
 reply. 
 
10 
 
 u 
 u 
 
 a: 
 
 O 
 
11 
 
 The reply from the transponder is also 
 fed to the modulator IC2 at pin 9. R32 
 and R34 set the f^ and f^ frequencies 
 while R30 sets the output level as mea- 
 sured at TP6. When IC4 times out, DIO 
 
 goes back on and IC3 goes off. With IC3 
 off, the output line becomes high imped- 
 anced and the next RCCM that transmits is 
 not loaded by the output impedance of 
 IC2, reflected through T2. 
 
 OTHER APPLICATIONS 
 
 As mentioned previously, the circuits 
 presented here are those of the original 
 development, which rely on the synchro- 
 nizing pulse from the transponder for 
 proper operation. These transponders are 
 located in the same enclosure as the 
 RCCM, and the synchronizing pulse has a 
 short distance to travel. If the trans- 
 ponders were to be located at a further 
 point down the line, the pulse (about 20 
 s long) would be dampened out by cable 
 capacitance. 
 
 Another application has been encoun- 
 tered that could make use of the modem 
 network. But, in this network, the 
 transponders would be located at dis- 
 tances up to 8,000 ft from the RCCM. 
 Operation of the circuits could not rely 
 on the synchronizing pulse. The RCCM 
 must gate the carrier on at the beginning 
 of the reply and then delay the reply 
 until the line stabilizes. This requires 
 the addition of a clock circuit and data 
 shift registers but provides operation 
 with a typical asynchronous serial chan- 
 nel format. 
 
 The main concept of the network still 
 remains the same. The LSM receives two 
 data words from the monitoring system (or 
 any serial computer channel) and returns 
 two data words in reply. The words must 
 be back to back at 4,800 Bd, with a posi- 
 tive start bit into the LSM and a nega- 
 tive into the RCCM. 
 
 A breadboard version of the modified 
 RCCM and LSM has shown successful re- 
 sults. Although the modification is dis- 
 cussed here, full documentation cannot be 
 available until field testing is com- 
 plete. For more information concerning 
 this modification, consult the author. 
 
 Refer to the RCCM schematic (fig. 8) 
 for a brief discussion of how the RCCM is 
 
 modified for testing of the breadboard 
 version. The reply from the transponder 
 enters the circuit through diode D9. The 
 leading edge of the reply toggles a flip- 
 flop that gates the carrier on, via IC3 , 
 and enables clocking of a 4-bit shift 
 register and a counter. A clock circuit, 
 identical to that of the LSM, is divided 
 down by the Q9 output of the counter and 
 fed to the clock input of the shift reg- 
 ister. The reply is then clocked through 
 the registers and arrives at IC2 , 3.5 bit 
 times later. A Q15 output of the counter 
 resets the flip-flop and thus resets the 
 counter and the shift register, and turns 
 the carrier off. 
 
 The modifications to the LSM can be 
 performed more simply. Refer to the LSM 
 schematic (fig. 6) and the timing diagram 
 (fig. 5). TP6 is the demodulated data 
 received from the RCCM. In the modified 
 version, the intermittent pulse at bit 
 time 10.5 will now occur at bit time 21.5 
 (the moment the transponder begin 's its 
 reply and turns on the carrier) , and the 
 reply will arrive at bit time 25 (3.5-bit 
 delay from the shift register of the 
 RCCM). The LSM can be modified to open 
 the channel at bit time 24, thereby miss- 
 ing the intermittent pulse and enabling 
 the reply. The channel can then be 
 closed 24 bit times later using the ex- 
 isting 24-bit timer. 
 
 To open the channel, the pulse of wave- 
 form 3 is moved to bit time 24 by AND'ing 
 the Q13 and Q14 outputs of IC3. The re- 
 sultant pulse can then be fed to IC7 pin 
 2, ICg pin 6, and IC5 pin 8. The output 
 of IC7 pin 11 should be disconnected, and 
 the time-out circuit of IC13 is no longer 
 needed. 
 
12 
 
 CONCLUSION 
 
 The modem network described in this re- 
 port is an inexpensive modulation tech- 
 nique for communication systems operating 
 in a polling mode. As such, it can com- 
 municate to an unlimited number of re- 
 motes at a relatively high baud rate. 
 
 The circuitry presented can be easily 
 modified to accommodate different baud 
 rates and word lengths. The procedures 
 here and in the indicated references can 
 be used to calculate component values for 
 the modulator and demodulator circuits. 
 The filter circuit is more complicated 
 and may not be necessary for many applic- 
 ations. If independent lines are used 
 for transmit and receive, and if circuit 
 crosstalk and noise are reasonably low, 
 then the demodulator will operate with a 
 reasonably low bit-error rate. 
 
 With an output power level of dBm, 
 the dynamic range of signal detection is 
 
 50 dB. Laboratory tests show that with a 
 signal-to-noise ratio (SNR) of 20 dB , the 
 error performance is on the order of 
 10"^. Preliminary field testing with 
 five remote modems distributed along the 
 north and south sections of the mine 
 revealed line losses averaging 6 dB/mi 
 and error rates about 10"^. If the 6-dB 
 loss per mile is typical, distances be- 
 yond 5 miles would degrade the perform- 
 ance and result in error rates greater 
 than 10" 5. 
 
 The effects of impulse noise induced 
 into the communication lines from load 
 switching or other sources could also re- 
 duce the SNR enough to increase error 
 rates. As in any modulation scheme, the 
 circuit is never assumed free from error 
 and some type of information protocol 
 should be considered. 
 
13 
 
 APPENDIX A.— FREQUENCY SELECTION AND FILTER DESIGN 
 
 FSK signals are commonly used to trans- 
 mit digital information over telephone 
 lines. In this type of modulation, the 
 carrier signal is shifted between two 
 discrete frequencies to encode the binary 
 data. These two frequencies are produced 
 by a function generator that is keyed 
 with the binary data. This generator is 
 then referred to as the modulator. A de- 
 modulator, such as those of phase-locked 
 loop (PLL) circuits, is then centered be- 
 tween the two frequencies and reproduces 
 the binary signal as the frequencies 
 shift back and forth. 
 
 The specifications of the function gen- 
 erators and PLL circuits include guide- 
 lines for selecting the two discrete fre- 
 quencies. These guidelines are based 
 upon the baud rate and carrier frequency, 
 which are determined by the available 
 bandwidth of the communication channel. 
 
 The typical communication channel for a 
 monitoring system utilizes telephone wir- 
 ing. Twisted-pair conductors of No. 16 
 or No. 18 gauge are very common sources 
 of mine wiring. In an environment free 
 from electromagnetic inteference, this 
 type of wiring could carry 4,800 Bd com- 
 munications over a 50-kHz carrier for 
 many miles. But documented evidence^ 
 shows that electromagnetic interference 
 in average mines is significant enough to 
 limit this kind of coimnunication to a 
 mile or two. 
 
 Shielding twisted-pair wiring improves 
 the immunity to electromagnetic interfer- 
 ence but increases the overall cable ca- 
 pacitance. This increase may be more 
 detrimental to a 50-kHz carrier in the 
 way of line loss than would the interfer- 
 ence. The environment then must be con- 
 sidered when selecting a cable, the baud 
 rate, and the carrier frequency for a 
 communication channel. In designing sys- 
 tems for general purposes, severe or 
 worst cases should be assumed or a 
 
 ^Bredeson, J. G., J. L. Kohler, and H. 
 Singh. Data Security for In-Mine Trans- 
 mission. Final Report — Part I (contract 
 J0308024, Univ. OK). BuMines OFR 76-81, 
 1981, 99 pp.; NTIS PB 81-221988. 
 
 variety of communication channels should 
 be designed. 
 
 The function generator and PLL circuits 
 used in the mine monitoring system modem 
 network are the IC's XR2206 and XR2211, 
 respectively. The specific design using 
 these IC's is covered in appendix B. 
 However, several operational characteris- 
 tics of the PLL must be considered here 
 in first selecting the frequencies of op- 
 eration and filtering requirements. 
 
 Application note AN-Ol^ gives the fol- 
 lowing guidelines for calculating non- 
 standard frequencies: 
 
 • The lower frequency, f,, must be at 
 least 55% of the upper frequency, f2 
 (less than a 2:1 ratio). 
 
 • Select fi and f.2 higher than the 
 baud rate, fp, for minimum pulse width 
 jitter. 
 
 • For maximum fj and f2 spacing (where 
 the ratio is close to 2:1), use the rela- 
 tionship (f2 - fi)/fp > 83%. 
 
 • For narrower f^ and f2 spacing, use 
 the relationship (f2 - f])/fp > 67%. 
 
 Athough "narrow spacing" is not too 
 well defined, some worked-out examples of 
 the application note for standard modems 
 result in the following frequency ratios 
 and the relationships used: 
 
 Baud 
 rate 
 
 Frequency 
 ratio 
 
 Relationship, 
 
 % 
 
 300 
 
 300 
 
 1,200 
 
 1.10:1 
 1.18:1 
 1.82:1 
 
 67 
 67 
 83 
 
 The communication channel at the Lu- 
 cerne No. 8 Mine would make use of two 
 available pairs in a multipair telephone 
 cable. Most of the pairs in the bundle 
 are used for telephone communications to 
 the underground section. For this rea- 
 son, the frequencies selected for modem 
 operation should be above the baseband of 
 voice signals. Since the pairs available 
 for use are direct wires (no telephone 
 loading coils or other circuitry), the 
 
 ^Exar Integrated Systems, Inc. (Sunny- 
 vale, CA) . Application Data Book. June 
 1981 , 80 pp. 
 
14 
 
 upper limit on the frequencies could be 
 based on propagation losses. 
 
 In the frequency spectrum (see figure 
 1)-^ for the available communication chan- 
 nel, f ] and the general shape along the 
 bandpass curve to its left should be 
 above the voice band, and f^ should be as 
 low as possible to minimize line losses 
 and standing waves that might occur as a 
 result of mismatching: preferably below 
 50 kHz. 
 
 From the guidelines and worked-out ex- 
 amples given above, the consideration 
 that the carrier frequencies may lie 
 around the 20-kHz range, and the 4,800-Bd 
 rate of the monitoring system, a rela- 
 tionship of 75% would work well. 
 Therefore, 
 
 (f2 - f ,)/fp = 0.75. (A-1) 
 
 Then the frequency spacing, df , is 
 
 nine resistors. The general layout of 
 the circuit is that of figure A-1. This 
 circuit is used with the demodulators for 
 group 1 (fi and £2) and group 2 (f^ and 
 f4). Using the monograph in the applica- 
 tion note, a bandstop-to-bandpass ratio 
 of 2:1 for this filter would yield a min- 
 imum bandstop attenuation of 24 dB. This 
 will be sufficient attenuation for out- 
 of-band noise and for crosstalk inter- 
 ference between the two groups of 
 frequencies. 
 
 This application note also recommends 
 that the filter bandpass be twice the 
 mark-space separation. Then the band- 
 pass, bp, from equation A-2 is 
 
 bp = 2 df = 2 * 3,600 Hz 
 
 = 7,200 Hz, (A-3) 
 
 and the bandstop, bs , is 
 
 df = f2 - f 1 = 0.75 fp = 0.75 * 4,800 
 
 bs = 2 bp = 2 * 7,200 Hz 
 
 = 3,600 Hz. 
 
 (A-2) 
 
 = 14.4 kHz. 
 
 (A-4) 
 
 Application note AN-03^ suggests the 
 use of a three pole-pair Chebyshev filter 
 with a bandpass ripple of 1 dB. The jus- 
 tification is that the filter can be con- 
 structed with just three operational am- 
 plifiers (one IC), six capacitors, and 
 
 o 
 
 -"Figure numbers without an A- prefix 
 refer to figures in the main text. 
 '*Work cited in footnote 2. 
 
 One further consideration before the 
 frequencies can be determined is that of 
 harmonic interference between the two 
 groups. The function generators used in 
 the modulator have a sine wave output 
 with low distortion. When the frequen- 
 cies are selected, the third harmonic of 
 frequencies in the first group must not 
 lie in the spectrum of the second group. 
 More precisely, the third harmonic of the 
 
 R1A 
 
 o-wvMHh^ 
 
 R2A 
 
 V <7 
 
 V V 
 
 FIGURE A-1. • Chebyshev bandpass filter. 
 
15 
 
 lowest frequency in the bandpass of the 
 first group is greater than the highest 
 frequency in the bands top of the second 
 group. Translated into equations, it is 
 
 3(fcl2 - bp/2) > fc34 + bs/2, (A-5) 
 
 where fcl2 = carrier frequency of 
 group 1, 
 
 and fc34 = carrier frequency of 
 group 2. 
 
 Substituting equations A-3 and A-4 into 
 A-5 and solving for the carrier of the 
 first group yields 
 
 fcl2 = fc34/3 + 6,000 Hz. (A-6) 
 
 The closest the two groups could be is 
 the point where the highest frequency in 
 the bandstop of the first group is equal 
 to the lowest frequency in the bandpass 
 of the second group, or 
 
 fcl2 + bs/2 = fc34 - bp/2. (A-7) 
 
 Then, from A-3 and A-4, 
 
 fc34 = fcl2 + 10.8 kHz. (A-8) 
 
 The simultaneous solutions to A-6 and A-8 
 yield 
 
 fcl2 = 14.4 kHz (A-9) 
 
 and fc34 = 25.2 kHz. (A-10) 
 
 Then, from A-2, 
 
 f , = fcl2 - df/2 = 14.4 kHz 
 
 - 3,600 Hz/2 = 12.6 kHz, (A-U) 
 f2 = fcl2 + df/2 = 14.4 kHz 
 
 + 3,600 Hz/2 = 16.2 kHz, (A-12) 
 fj = fc34 - df/2 = 25.2 kHz 
 
 - 3,600 Hz/2 = 23.4 kHz, (A-13) 
 
 and 
 
 f4 = fc34 + df/2 = 25.2 kHz 
 
 + 3,600 Hz/2 = 27.0 kHz. (A-14) 
 
 The application note AN-025 contains a 
 worked-out example for the design of the 
 filter shown in figure A-1. An addition- 
 al reference may be necessary for better 
 understanding of filters as used in oper- 
 ational amplifiers. In particular, an- 
 other reference^ was consulted for better 
 definition of the individual component 
 calculations. All of the capacitors of 
 figure A-1 can be set equal, and the re- 
 sistor values become 
 
 Rlx 
 
 Qx 
 
 Ho 2Trfx C 
 
 R2x = 
 
 R3x = 
 
 Qx 
 
 (2Qx - Ho) 2Trfx C 
 2Qx 
 
 2Trfx C 
 
 where Qx = the Q (ratio of reactance to 
 resistance) of the individ- 
 ual stages A, B, and C, 
 
 fx = the center frequency 
 of the stage. 
 
 Ho = the gain of the stage, 
 
 and C = the value of capacitance 
 chosen. 
 Either of the references (cited in 
 footnotes 2 and 6) can be consulted for 
 the procedures to determine the Q and 
 center frequencies of the stages. The 
 gain is a free parameter and can be cho- 
 sen for the desired amplitude response. 
 In practice, Rl is much greater than R2; 
 
 -"Work cited in footnote 2. 
 
 ^Graeme, J. G. , G. E. Tobey, and L. P. 
 Huelsman. Active Filters. Ch. 8 in Op- 
 erational Amplifiers, Design and Applica- 
 tions. McGraw-Hill, 1971, p. 293. 
 
16 
 
 so R2 can be used to trim the Q. Then, 
 to adjust the center frequency, R2 and R3 
 can be simultaneously adjusted by the 
 same percentage with negligible effect on 
 the Q. 
 
 Figure A-2 is a set of reproduced pic- 
 tures from a spectrum analyzer, showing 
 the response of the filter used in the 
 LSM. This filter is centered at 25.2 
 kHz, the carrier of the RCCM, and has an 
 overall Q of 3.5 with unity gain in the 
 bandpass. Amplitude is given in decibels 
 with a reference level of 1 V root mean 
 square (RMS). 
 
 Panel 1 , of figure A-2 shows a noise 
 source used as an input to test the indi- 
 vidual stages of the filter. Panel 2 
 shows the output of stage A with the 
 above noise at the input. This stage has 
 
 a Q of 14 and a gain of 12 dB (Ho = 4) at 
 its peak frequency. Panel 3 shows the 
 output of stage B with noise at the in- 
 put. It is a reflection of stage A, 
 about the geometric center of the filter, 
 with the same characteristics. Panel 4 
 shows the output of B with noise at the 
 input of A and the output of A into B. 
 Panel 5 shows the output of stage C with 
 noise at the input. This stage has a Q 
 of 7 and a gain of 12 dB at 24.94 kHz, 
 the geometric center of the 3-dB bandpass 
 points. Panel 6 shows the final output 
 of stage C with noise at the input of A, 
 and Panel 7 shows a comparison of input 
 to output. 
 
 Figure A-3 is a reproduced picture of 
 the spectrum response from an RCCM. The 
 
 -40 
 
 -10 
 
 -20 
 
 -30 
 
 -40 
 
 -30 
 
 -40 
 
 -50 
 
 -60 
 
 . .y^ ^'''''''''^. . - 
 
 28.75 
 
 25.25 
 
 -10 
 
 +2 +4 +6 -t-8 +10 
 
 -6 -4 -2 +2 +4 +6 +8 +10 -|0 -i 
 
 FREQUENCY, kHz 
 
 FIGURE A-2. - Filter response. ], Noise source input; ~, stage A; 3, stage B; J, stages A 
 combined; .5, stage C; 6, stages A, B, and C combined; 7, input-to-output comparison. (Number 
 each panel is the center frequency of the panel, in kilohertz.) 
 
 and B 
 below 
 
17 
 
 top waveform was measured at TP4 of the 
 LSM (fig. 6) and shows the unfiltered re- 
 sponse. The two peaks correspond to mark 
 
 and space frequencies transmitted from 
 the RCCM. The bottom trace was measured 
 at TP5 and shows the filtered response. 
 
 -70 
 
 TP4 raw 
 
 15 17 19 21 23 25 27 
 
 FREQUENCY, kHz 
 
 FIGURE A-3. - RCCM filtered response. 
 
 29 
 
 33 
 
 35 
 
18 
 
 APPENDIX B. — MODULATION DESIGN AND CALIBRATION 
 
 The modulator and demodulator used for 
 the LSM and RCCM IG's are the XR2206 and 
 XR2211, respectively, from Exar Integrat- 
 ed Systems, Inc. The application notes^ 
 on these circuits cover the theory of op- 
 eration. The justifications for the 
 equations as well as worked-out examples 
 are included in those notes. 
 
 The design solutions for the RCCM and 
 LSM demodulators are presented first. 
 Refer to figure B-1. 
 
 Choose the value of the timing resistor 
 (RO) between 10 and 100 kQ or -50 kf2. 
 
 Then CO = 1/(R0 * fo) = 0.00139 pF; 
 
 let CO = 0.001 uF, 5% for 
 convenience, 
 
 RO = 1/(C0 * fo) = 69.4 kQ 
 = 64.9 k$7, 1% 
 
 + 10 kO, potentiometer. 
 
 RCCM DEMODULATOR 
 
 Given that f, = 12,600 Hz and £2 = 
 16,200 Hz, calculate PLL center frequency 
 as fo = (f , + f2)/2 = 14,400 Hz. 
 
 Exar Integrated Systems, Inc. (Sunny- 
 
 vale, CA). Function Generator Data Book. 
 May 1981, 53 pp.; Phase-Locked Loop Data 
 Book. Feb. 1981, 85 pp. 
 
 Calculate Rl to give a lock range equal 
 to the mark-space deviation as 
 
 Rl = (RO * fo)/(f2 - fl) 
 = 277.7 kfl -270 kfl. 
 
 Calculate CI to set loop damping equal 
 to 0.5 as 
 
 FSK 
 input 
 
 iCD 
 
 VCO 
 fine tune 
 
 FIGURE B-1. - Demodular circuit. 
 
19 
 
 CI = CO/4 = (0.001 uF)/4 
 
 = 250 pF -270 pF. 
 
 Calculate CF for data filer time con- 
 stant as 
 
 CF(uF) = 3/baud rate = 3/4,800 
 
 = 625 pF -620 pF. 
 
 The capture range, dfc, is recommended 
 to be 80% to 95% of the lock range. 
 Picking 90% yields dfc = 3,240 Hz. For 
 the full-tracking bandwidth, the lock- 
 detect filter capacitance is given as 
 16/2 dfc (in microfarads). 
 
 CD (yF) = 16/(2 * 3,240) = 0.0025 pF. 
 
 LSM DEMODULATOR 
 
 Given that fj = 23,400 Hz and f^ = 
 27,000 Hz, calculate PLL center frequency 
 as fo = (f3 + f4)/2 = 25,200 Hz. 
 
 The calculated values follow, as for 
 the RCCM demodulator: 
 
 CO = 1/(R0 * fg) = 0.00079 yF. 
 
 Let CO = 0.001 uF, 5% for 
 convenience. 
 
 Then RO = 1/(C0 * fg) = 39.7 kJ2 
 = 34.8 k^, 1% + 10 ka 
 potentiometer, 
 
 Rl = (RO * fo)/(f2 - fl) 
 = 277.7 kD. -270 kn, 
 
 CI = CO/4 = (0.001 yF)/4 
 = 250 pF -270 pF, 
 
 CF = 3/baud rate = 3/4,800 
 = 625 pF -620 pF, 
 
 CD = 16/(2 * 3,240) 
 = 0.0025 yF. 
 Next, the design solutions for the RCCM 
 and LSM modulators are presented. Refer 
 to figure B-2. 
 
 0>2V p-f| 
 
 ^y . Q<IV-'fo 
 
 Keying Y ^ 
 
 input -L 
 
 X 
 
 FIGURE B-2. - Modulator circuit. 
 
20 
 
 RCCM MODULATOR 
 
 RCCM CALIBRATION 
 
 The RCCM must transmit the mark and 
 space frequencies of the LSM demodulator, 
 so fj = 23,400 Hz, f4 = 27,000 Hz, and fg 
 = 25,200 Hz. 
 
 Typical values of Rl and R2 are -50 kO. 
 so CO is chosen from 
 
 CO = 1/(R * fo) = 1/(50 kn * 25,200) 
 
 = 794 pF -820 pF, 5%. 
 
 Then Rl and R2 are determined from 
 R = 1/(C0 * fo): 
 
 Rl = 1/(820 pF * 23,400) 
 = 52.1 kn = 46.4 kJ2, 1% 
 + 10 kfi potentiometer. 
 
 R2 = 1/(820 pF * 27,000) 
 = 45.2 kfi = 40.2 kQ, 1% 
 + 10 k^ potentiometer. 
 
 LSM MODULATOR 
 
 The LSM must transmit the mark and 
 space frequencies of the RCCM demodula- 
 tor, so f , = 12,600 Hz, f2 = 16,200 Hz, 
 and fo = 14,400 Hz. 
 
 As in the RCCM modulator, 
 
 CO = 1/(50 kfi * 14,400) 
 
 = 1,389 pF -1500 pF, 5%, 
 
 Rl = 1/(1,500 pF * 12,600) = 52.9 kO. 
 = 47.5 kn, 1% + 10 kQ 
 potentiometer, 
 
 R2 = 1/(1,500 pF * 16,200) = 41.2 kn 
 = 36.5 kfi, 1% + 10 kj^ 
 potentiometer. 
 
 Since the operation of this modem net- 
 work requires only one LSM, the con- 
 struction and calibration processes were 
 performed at the engineer's level. 
 Approximately 20 RCCM's were anticipated 
 for the Lucerne No. 8 monitoring system; 
 therefore, the construction and calibr- 
 ation process was clearly defined such 
 that the circuits could be built at the 
 technical level. 
 
 To aid in calibration of the RCCM, the 
 circuit of figure B-3 was developed and 
 is referred to as the RCCM calibrator. 
 This circuit is adjusted to transmit the 
 f 1 and f2 frequencies of the LSM and has 
 connectors that directly mate with the 
 RCCM. An XR2207 function generator is 
 used to modulate the XR2206 with a 1-kHz 
 and a 2.4-kHz square wave. The 1 kHz 
 serves as a convenient scale when check- 
 ing demodulated data symmetry on an os- 
 cilloscope, and the 2.4 kHz checks the 
 demodulator's performance at an equiva- 
 lent 4,800-Bd rate. 
 
 The calibration procedure sets the fre- 
 quencies and power levels of the RCCM. 
 Although the RCCM may be calibrated imme- 
 diately after assembly, it is recommended 
 that a final calibration be performed 
 after a 168-h burn-in period. 
 
 Test equipment 
 
 • Oscilloscope, single trace 
 
 • RMS voltmeter, 600 Q 
 
 • Frequency counter 
 
 • RCCM calibrator 
 
 Procedure 
 
 1. Connect plugs from calibrator to 
 RCCM and power outlet. Set SI to f^ pos- 
 ition and S2 to B,. The CD and gate ind- 
 icators should be on. 
 
 2. Connect RMS voltmeter and frequency 
 counter in parallel. Attach their com- 
 mon, along with the scope common, to 
 TP7. 
 
 3. Connect voltmeter-counter probe to 
 TPl. The voltmeter should read about 
 -10 dB. 
 
 4. Move the voltmeter-counter probe 
 to TP2. The voltmeter should read about 
 -6 dB. 
 
 5. Connect scope probe to TP3. Set 
 vertical scale to 5 V per division and 
 the horizontal scale to 0.5 ms , and syn- 
 chronize the trace if possible. 
 
 6. Adjust balance control for equal 
 positive and negative durations. Depress 
 S2 to B2 position. The scope trace 
 should remain with equal positive and 
 negative durations at a higher frequency 
 (a small amount of jitter may be noted). 
 
 7. Connect voltmeter-counter to TP5. 
 
21 
 
 r 
 
 o.ovfT 
 
 C5 
 
 RI4 
 82 
 
 RI5 
 25 
 
 kxi 
 
 /J 
 
 J // 
 
 D 
 
 10 
 
 6 8 
 
 9 12 
 
 Rll 
 4.7 kfl 
 
 RI2 
 3.9kn ^ 
 
 
 
 
 
 RI8^ 
 
 ka ' 
 
 > 
 > 
 
 > 
 
 J1 -2 
 J1-4 
 
 ^si, 
 
 1 
 
 
 ii 
 
 J1-1 
 
 
 
 
 Ji -3 
 
 . ^ 25 ^ 
 
 B, '^^ 
 
 cw 
 
 C6 
 I/.F 
 
 S2 
 
 t — v^AA- 
 
 RI3 
 ii 4.7 kil 
 
 B' 
 
 I 
 
 15-V 
 power 
 supply 
 
 i- C7 
 lO/xF 
 
 ac 
 power 
 
 FIGURE B-3. - RCCM calibrator schematic. 
 
22 
 
 8. Adjust level for dB on 
 voltmeter. 
 
 9. Adjust £4 for 27.00 kHz. 
 
 10. Move voltmeter-counter probe to 
 TP6. There should be no output (less 
 than -50 dB). 
 
 11. Switch SI to fj position. The 
 gate indicator should go out. 
 
 12. Readjust level for dB on volt- 
 meter (you should have to increase it by 
 about 5 dB). 
 
 13. Adjust fj for 23.40 kHz. 
 
 14. Calibration is complete. Remove 
 test equipment and seal RCCM adjustments. 
 
 <^^ 
 
 FIGURE B-4, - LSM printed circuit layout. 
 
23 
 
 LSM CALIBRATION 
 
 The modulator transmit frequencies of 
 the LSM are adjusted while controlling 
 the digital input at J2 pin 2 (see figure 
 6). 2 The output level, as measured at 
 TP3, is adjusted for dB with a 600-i^ 
 resistor connected across TBl-4 and TBl- 
 5 The reference oscillator is adjusted 
 while monitoring at TP9 and the level set 
 
 to -50 dB. . . 
 
 To calibrate the demodulator, it is 
 best to have a freshly calibrated RCCM. 
 Modulate the RCCM, at TP4, with a 1-kHz 
 square wave and connect its output to the 
 input of the LSM. Adjust the demodulator 
 for symmetry as measured at TP6. 
 
 Figure B-4 is the printed circuit lay- 
 out of the LSM shown in figure 6. Figure 
 B-5 is the layout for the LSM line split- 
 ter of figure 7. And lastly, figure B-6 
 is the layout of the RCCM of figure 8. 
 
 2Figure numbers without a B- prefix re- 
 fer to figures in the main text. 
 
 FIGURE B-5. - Line splitterprintedcircuit layout. 
 
 FIGURE B-6. - RCCM printed circuit layout. 
 
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