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MICROCOPY RESOLUTION TEST CHART
NATIONAL BUREAU OF STANDARDS - 1963
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conf.670520 --
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9.21-67
D21-67
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MASTER
A LOW-NOISE WIDE-BAND CURRENT PREAMPLIFIER
FOR USE WITH SEMICONDUCTOR DETECTORS
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J. K. Millard and T. V. Blalock
Oak Ridge National Laboratory
Oak Ridge, Tennessee
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Presented at the Semiconductor Nuclear Particle Detectors and circuits
Conference, May. 15-18, 1967, Gatlinburg, Tennessee..
SC
Research sponsored by the U. S. Atomic Energy Commission under
contract with the Union Carbide Corporation.
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DISTRICTION OF THIS DOCUMENT, ES UND MATED
·
A LOW-NOISE WIDE-BAND CURRENT PREAMPLIFIER
FOR USE WITH SEMICONDUCTOR DETECTORS
,
J. K. Millard and T. V. Blalock
Oak Ridge National Laboratory
Oak Ridge, Tennessee
Introduction
A new low-noise wide-band current preamplifier was developed for use in
fast-signal processing systems for semiconductor detectors. This preampli-
fier consists of a single current-gain section with paralleled field-effect ...
transistors in the input stage. The equivalent white-noise current of the
preamplifier 18 about an order of magnitude below that of previously reported;
current preamplifiers of comparable bandwidth. The current gain and 3 db
bandwidth of the preamplifier are about 500 and 50 MHz, respectively.
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Noise Characteristics of Current Preamplifiers
The current preamplifier configuration (Fig. 1) developed by Millard
...i can accommodate either field-effect transistor (FET) or bipolar transistor ,
Input stages. The dominant noise components in this configuration are a ...:
white component and oa jhigh-frequency component that can be represented by an ...
equivalent noise resistance Rovshunting the preamplifier input. For the bi-..
polar transistor inputs the noise resistance can be determined with the help .
of the hybrid-pi nolse" modeland analytical techniques like those outlined
by Kennedy." The noise resistance in the midband region of the preamplifier
(Fig. 1) signal transfer function A(w) 18 approximately
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. Research sponsored by the U. S. Atomic Energy Commision under contract
with the Union Carbide Corporation.
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ORNL-DWG 57 - 4803R
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Mg. 1. Current preamplifiers: (a) bipolar transistor input, (D) field-effect
transistor input.
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É
ET
[W
] [2+ (own
where I, is in m, R, in ohns and w in radians per sec. The parameters
and We are the base spreading resistance (ahms) and the common-eadtter
: current gain respectively of the inputi.transistor.
For an FET input stage (Fig. 1b) .
*** TCE 600 + C * orna
where Ry 18 the equivalent white noise resistance of the preamplifier (ex-
cluding the thermal noj.se contribution of Rp), Ca 18 the FET gate-to-source
capacitance, and Cod is the FET gate-to-drain capacitance. The C2/3 term
in Eq. (2) accounts for the induced gate noise.“ If the transconductance &
of the FET 1s sufficiently high (as in the case of paralleled FETS), R. 18
approximately 0.7/set.
The power spectrum of the preamplifier output noise 18 given ayrroxi-
mately by
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W(w) Je
(3) si
is
Since the magnitude of the signal transfer function (AC) | MU decrease
more rapidly than 6 db per octave for frequencies beyond the high frequency
3 dd point, the general form of the power spectrum predicted by Eq. (3) 18
as shown in Fig. 2. The spectrum 18 composed of a waite component and a fre-
quency dependent component that increases to a peak value in the vicinity
of the high frequency 3 db point.
Minimum white noise for either bipolar transistor or FET input occurs v
: when the white noise resistance Rome Rp; that 18, when the preamplifier
ORNL- DWG 67-7
:: Nlw)- OUTPUT NOISE POWER
.. (mean-squared volts or amp)
HIGH FREQUENCY
NOISE CORNER -
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-HIGH FREQUENCY 3 db POINT
FOR THE SIGNAL
TRANSFER FUNCTION
FPEQUENCY (Hz)
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Fig. 2. Power spectrum of the output noise of a current preamplifier
using either a bipolar or field-effect transistor input stage,
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white noise contains no contribution from the active devices. This case 16
practically impossible to achieve with presently availadle bipolar transie-
tors because the required low value of emitter current I, produces a re-
IS
white noise contribution of the · FET 18 easily made negligible, since R can ...
be much less than R, in Eq. (2), even for the case of a large low-frequency
noise component in R. Tlus, for the same value of feedback resistor Rp, the
white noise of a preamplifier with an FET Input 18 generally much less than
one with a bipolar transistor input.
Inspection of Eqs. (1) and (2) reveals that the high frequency noise of
the FET input would usually de higher than that of the bipolar laput near the
high frequency 3 db polat. Also the high-frequency noise corner Ion (waste
component power equals high-frequency component power) is usually much lower
for the FET input. The greater noise at high frequencies for the FET input
could be a significaut disadvantage 11 the output current pulse were used to
trigger a fast discriminator. The much lower value of white noise would,
however, be an important advantage 1f the current puilse were used in a time-
Invariant pulse-height analyzing system.
The equivalent noise current spectrum of the two types of current.pro
amplifiers are compared in Fig. 3. The emitter current of the input stage of
the bipolar-Input preamplifier was set about as low da was practical (0.5 ma)
for reasonable gain-bandwidth. The input device of the PET preamplifier was
assumed to have a transconductance of 20,000 umho. The noise of the YHT 10-
put preamplifier 18 much lower than that of the bipolar Input proamplifier for ::
frequencies below 10 M2, which 18 the frequency domaia where most pulse-height
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analyzing systems operate.
A Low-Noise Wide-Bard Current Preamplifier with an FET Input stage.
When an FET 18 used as the first stage of the current preamplifier
(Fig. 1b) minimum noise operation requires a high value of Rp (EQ. 3).
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ORNL-DWG 67-7817
#AAAAA#A
* BIPOLAR TRANSISTOR INPUT - Cin = 20 pf, le=0.5 ma,
In=25 St, Bo=50, Ry=50k, ff* 50 MHz
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EQUIVALENT NOISE CURRENT
IH
3.3 MHZ
1496 MHZ
10
HITTITTI
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FET INPUT - Cin=20 pf, Cgs = 14 pf, Cgd = 3.2 pf,
Rn=35 S2, Ry =50k, fy 50 MHZ I L
104.
105 .. 406
FREQUENCY (Hz)
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cas.

resistance. .
Fig. 3. Equivalent noise current for current preamplifiers with a high value of feedback
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However, wide bandwidth, which 1s necessary for amplification of last current.
pulses from semiconductor detectors, requires a low value of Rp. The 10-to-
90% rise time for the preamplifier of Fig. 1b is given approximately by. '
where K 18 a constant, Car is the total capacitance loading the input node,
Co is the total capacitance loading the output of the shunt-feedbaca section,
and Em is the transconductance of the FET. The expression indicates that the
rise time increases directly as the square root of the feedback resistance.
. The constant K depends on the system damping factor which is defined
by
Co + 204 + Bracele
27 Cat Co Reborn
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A 6 of about 0.7 with allow the shortest rise time without excessive
overshoot. The value of K associated with this 6 is about 2.2. With a 50 :
kliohm feedback resistor, the capacitance ce across the resistor (approxi-
mately 0.2 pp for a 1/4 w resistor) will give a reasonable value of 6. I ::
Co becomes too large, the system will become overdamped and the pole deter-
mined by the RoCe product will dominate the closed loop response.
The capacitance loading of the input due to the detector and associated.
stray capacitance is included in the value of C, and could produce an apprea :
ciable rise-time limitation; however, parallel FETs in the input minimizes
this effect. With parallel FETS, the pert of Cat due to the preamplifter is
Increased so that the detector and stray capacitances become a smaller part
of the total Input capacitance. As can be seen in Eq. (4), the increase in
Cat due to the added FETS 18 offset by a proportional increase in &nThus
parallel FETs at the luput Increases the gain-bandwidth product of the
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preamplifier when the detector capacitance is an appreciable portion of the
total input capacitance. With four 2N4417 FITs operating in parallel (total
Bm 20,000 kimhos), the preamplifier had a rise tine of approximately 5 nsec
with a feedback resistance of 10 kilobiria and with a peaking capacitor added.
to the emitter of the output stage.
. To allow further increases in Rp with no Increase in rise time, more
gain was needed in the shunt-feedback loop. This additional gain was real-
ized by inserting a wide-band bipolar transistor current-gain section" be-
tween the FET drain and the emitter of the common-base stage (Fig. 4). This
two-transistor gain section has a very low input Impedance and a relatively:
high output Impedance. For wide bandwidth, both of these conditions are
Ideal for an internal gain stage at this position in the shunt-feedback loop.
. With a 2000-Maz transistor for 23, the current gain of this internal stage
can be set as high as 10 without producing stability problems in the preampli. .
fier.
With an R. value of 51 kilobms, the preamplifier of Fig. 4 had a measured .
current gain of 513 when loaded with 93 ohms. The rise time was 7.5 ngec with
about 10% overshoot, and could be decreased to 5 nsec by adding a peaking ca-
pacitor to the emitter of Q6. The measured equivalent parallel white noise
resistance was 49 kilohms (ENI = 0.58 pa/Hz*). The high frequency noise corner
was 2.7 MHz for Cen = 20 pf. The peaks high frequency ENI was about 3.9 pa/Hz.
at .a' frequency of approximately 35 MHz.: :
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Acknowledgements
We are grateful to C. H. Nowlin for his assistance in the computer analysis
of the preamplifier circuits and to E. J. Kennedy for discussion on the noise an-
alysis. We thank I. S. Sherman and R. G. Roddick of the Argonne National Labora-
tory for several very helpful discussions on the high frequency noise spectrum of
current preamplifiers.
ORNL-OWG 67-4804R
+ 24
100 uh
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2.37
: 2.376
{round
TRANSISTORS:
Q,-FOUR 2N4447 FET UNITS
. OPERATING IN PARALLEL
Q2, Q5, 06--2N3579
Q2-K2114 B (KMC CORP.)
Q4-2N4261
Qy-2N3933
ALL CAPACITORS IN UNLESS INDICATED
ALL RESISTORS W MF $ 1%
Q3
3 3.3k3100
287
CURRENT
INPUT 0.2
BNC crit
70.
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26k
0.2 pt
riba
CURRENT
OUTPUT
BNC
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51.1k
33.83k
34.64 k
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35.62k
5-25 pf *
5
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3kB
in 204
$100
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Fig. 4. Low-noise wide-band current preamplifier.
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References
"1. J. K. Millard, "New Shunt-Series Nanosecond Pulse Amplifier," Rev. Sci. 7
Instr. 38 No. 2, Feb., 1967.
2. R. Nutt, T. V. Blalock, and J. F. Pierce, A Transistor Hybrid-Pi Noise
Model, Scientific Report No. 3, ORNL Contract W7405 Eng. 26 (2160) Elect.
Engr. Depc., Univ. of Tenn. (Oct. 1, 1962).
:::
-...
3. E. J. Kennedy, A Study of the Theoretical and Practical Limitations of Low-
Current Amplification by Transistorized Current-Feedback DC Electrometers,
ORNL-IM-1726 (Ph.D. Thesis, Univ. of Tenn.), April, 1967. .
4. A. van der Ziel, "Gate Noise in Field-Erfect Transistors at Moderately
High Frequencies," Proc. IAEE 51, 461-67 (March, 1963).
5. C. J. Rush, "New Technique for Designing Fast Rise Transistor Pulse
Amplifiere," Rev. Sci. Instr. 35, No. 2, Feb., 1964..
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END

DATE FILMED
2 / 23 / 68
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