UNIVERSITY OF 
 
 ILLINOIS LIBRARY 
 
 AT URBANA-CHAMPAIGN 
 
uiucdcs-r-7U-677 
 hO 677 
 
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 HARDWARE DESIGN OF A REAL-TIME MUSICAL SYSTEM 
 
 by 
 
 Sergio Franco 
 
 October, 197 1 * 
 
uiucdcs-r-t 1 +-6tt 
 
 HARDWARE DESIGN OF A REAL-TIME MUSICAL SYSTEM 
 
 BY 
 SERGIO FRANCO 
 
 October, 197 1 * 
 
 Department of Computer Science 
 University of Illinois 
 Urbana, Illinois 6l801 
 
 This work was submitted in partial fulfillment of the requirements 
 for the degree of Doctor of Philsophy in Computer Science, at the 
 University of Illinois. 
 
Ill 
 
 HARDWARE DESIGN OF A REAL-TIME MUSICAL SYSTEM 
 
 Sergio Franco, Ph.D. 
 
 Department of Computer Science 
 
 University of Illinois at Urban a- Champaign, 197*+ 
 
 ABSTRACT 
 
 This thesis describes the hardware design of a hybrid system for the 
 composition and performance of electronic music in real-time. While analog 
 circuitry is primarily employed in the generation and processing of sounds, 
 digital circuitry is devoted to the exercise of control, under the immediate 
 supervision of the composer/performer. Such a digital vs. analog partition, 
 together with a proper choice of the man vs. machine interface, is intended 
 to satisfy the much emphasized musical need for the immediate, real-time 
 interaction between the composer /performer and his instrument. 
 
Digitized by the Internet Archive 
 in 2013 
 
 http://archive.org/details/hardwaredesignof677fran 
 
IV 
 
 ACKNOWLEDGMENTS 
 
 The author expresses his sincere gratitude to Prof. Michael Faiman, 
 his thesis advisor — and friend — for his indispensable support, advice and 
 suggestions throughout the development of this thesis. 
 
 The author is also pleased to credit Dr. J. Divilbiss with initiating 
 him into the fascinating field of electronic music. 
 
 This thesis has emerged from a two-year collaboration with composer 
 Salvatore Martirano and with the other members of the project. To his friend 
 Sal, the author expresses his warmest thanks for those endless, heated and 
 constructive discussions that have made this thesis possible. 
 
 The diligent typing and the beautiful drawings of this thesis have been 
 made by Ms. Evelyn Huxhold and under the supervision of Mr. Mark Goebel, 
 respectively, both of whom the author thanks for their precious collaboration, 
 
 This work is dedicated to Diana. 
 
TABLE OF CONTENTS 
 
 CHAPTER PAGE 
 
 I. INTRODUCTION 1 
 
 1.1 Importance of Real-Time in Electronic Music ------ l 
 
 1.2 Electronic Music Instruments ------------- 3 
 
 1.3 General Concept of New Instrument ----------- 6 
 
 l.U System Organization ------------------10 
 
 II. TONE SYNTHESIS 20 
 
 2.1 Harmonic Tone Generation --------------- 20 
 
 2.2 Pitch Control 25 
 
 2.3 Exponential Current Generation _--_-_______ 25 
 
 2.k Relaxation Oscillator -----------------30 
 
 2.5 Digital Waveshape Generation ------------- 3I4. 
 
 2.6 Programmable Mixer/Modulator ---------____ 38 
 
 III. ELECTRONICALLY PROGRAMMABLE FILTERS k2 
 
 IV. SOUND INTENSITY AND LOCATION 53 
 
 U.l Intensity and Location Control ------------ 53 
 
 k.2 Programmable Attenuator /Locator ------------ 5I+ 
 
 k.3 Envelope Generation ------------------ 6l 
 
 V. PERCUSSION ENSEMBLES 67 
 
 VI. SOUND DISTRIBUTION AND SWITCHING 72 
 
 6.1 Spatial Sound Distribution '-------— ------72 
 
 6.2 Audio Switching --------------------76 
 
 VII. CONTROL SYSTEM 82 
 
 7-1 Musical Parameter Control ---------------82 
 
 7.2 Programmable Binary Sequence Generation --------85 
 
 VIII. CONCLUSION 91 
 
VI 
 
 CHAPTER PAGE 
 
 LIST OF REFERENCES 93 
 
 VITA 95 
 
I. INTRODUCTION 
 
 1.1 Importance of Real-Time in Electronic Music . 
 
 With the advent of electronic music, the roles of the composer and the 
 performer have been placed in a new perspective. Traditionally, musicians 
 have tended to specialize in either one role or the other, or have tended to 
 play only one of the roles at a time. This has happened mainly because of 
 the considerable difficulty encountered when composing and performing are 
 carried on simultaneously. When a musician improvises at an instrument, he 
 is essentially composing music in real-time. This means he is thinking and 
 evaluating very quickly a series of possibilities out of which he makes 
 appropriate choices, which in turn lead to the execution of the various mechani- 
 cal motions necessary to produce the music he wants. Because of the difficul- 
 ties encountered in doing all these things at once, musicians have tended to 
 specialize either in the incept ion/ evaluation process (composition) or in the 
 execution process (performance), and have developed the notation of musical 
 scores to communicate among themselves. Thus, the composer need not be prin- 
 cipally concerned with the quick thinking and decision making of the impro- 
 visational process, but can pursue his inspection of musical possibilities on 
 his own time scale. What is even more important, to appreciate the effects 
 of his choices and changes, he does not need to try them out on actual instru- 
 ments because he can rely on his aural imagery, a faculty he has developed 
 through experience either by playing instruments or by listening to others 
 playing them, or both. 
 
 In developing his aural imagery, the musician is certainly helped a lot 
 by the fact that he deals with a limited set of instruments whose character- 
 istics extend over known and predictable ranges. In electronic music the 
 
situation is quite different. Because of the new kinds of sounds that elec- 
 tronic instruments are capable of synthesizing, it is very difficult for the 
 musician to develop the kind of musical imagery that may suffice to assist 
 him in composing on his own time scale and away from his instruments. If the 
 composer is to come up with something musically meaningful at all, it is 
 absolutely essential that he actually hear the effects of his choices, his 
 trials, and his alterations as he makes them, so that he can directly evaluate 
 them in the context of the whole composition. In other words, the concept of 
 direct feedback has come to play a dominant role in the electronic music com- 
 posing process. 
 
 Another important change brought about by electronic music involves the 
 relationship between performer and instrument, and is due to the much greater 
 detail in which the performer is required to control the various musical para- 
 meters. While in conventional instruments such parameters as timbres, attacks, 
 decays, etc. are, to a large extent, fixed, built-in features, in electronic 
 music they are left to the discretion of the performer, who must therefore 
 specify and control them directly. If this feature allows on the one hand 
 much more freedom of choice and experimentation, on the other it imposes a 
 more demanding control burden upon the performer, and it certainly renders 
 the communication among musicians more complex than with conventional instru- 
 ments, where the notation of the musical score is usually adequate. 
 
 As a result of the above discussion, it should be clear that in electronic 
 music the distinction between composer and performer ceases to exist, partly 
 because the composer needs to try out his musical ideas on the instrument by 
 himself, and partly because of difficulties of communication among different 
 people. Furthermore, having accepted the notion of direct feedback as an in- 
 dispensable ingredient of the improvisational process, it is of paramount 
 
importance that the composer/performer "be allowed to interact with his instru- 
 ment on the time scale of the music he is improvising, that is, in real-time. 
 
 1.2 Electronic Music Instruments . 
 
 The instruments available today to the electronic musician fall into one 
 of two categories: Analog music synthesizers and digital computers. 
 
 Analog synthesizers are self-contained systems consisting of a number of 
 basic building blocks which can be interconnected in a variety of configura- 
 tions to generate and modify sounds.* The basic sound-generating device is 
 the oscillator, which produces a predetermined number of waveforms, usually 
 the sinusoidal, pulse, saw-tooth and triangular waves. Starting with these 
 basic sounds, new waveforms can be obtained by means of mixers, filters, non- 
 linear signal processors, and other sound modifying devices. Another basic 
 sound modifier is the attenuator, which is used to control the dynamic charac- 
 teristics of the amplitude or loudness of sounds. 
 
 In order to allow musical parameter control, these circuits are designed 
 so that their characteristics can be altered externally by the operator, a 
 task normally accomplished by means of manual switches, potentiometers, piano- 
 like keyboards, patch cords, etc.** Recent advances in circuit technology and 
 design allow several circuit parameters to be electronically controllable by 
 
 * During the ' 6o's, the analog synthesizer scene was mostly dominated 
 by the instruments of R. A. Moog, operating on the East coast, and D. Buchla, 
 on the West coast. In recent years, encouraged by the enormous progress of 
 technology, other firms have joined in this venture, so that the market today 
 offers more than half a dozen lines of synthesizers, spanning a wide range of 
 quality, complexity, and price. Besides MOOG and BUCHLA, other synthesizers 
 widely known today are ARP and PUTNEY. 
 
 ** Because of the strong competition, most synthesizer manufacturers 
 neither publicize their designs nor provide circuit schematics with the pur- 
 chase of their equipment. To the author's knowledge, the only comprehensive 
 exposition of analog synthesis concepts and design techniques available to 
 date is offered in the papers by R. A. Moog (see references 2, 3). 
 
2 3 
 voltage signals. ' Hence, certain control responsibilities can be removed 
 
 from the operator and built into the system by having some circuits control 
 
 parameters of the others directly. In spite of this, a considerable amount 
 
 of manual operation is still required of the performer, mostly in the form 
 
 h 5 
 of knob turning and dial setting. ' 
 
 With lack of sophistication and flexibility in the control of the musical 
 parameters, the amount of influence the musician can exert upon his instrument 
 in real-time is rather limited. This broadening of the gap between amount of 
 control required and conditions for its implementation results in a further 
 aggravation of the basic dilemma of the improvisational process already dis- 
 cussed. 
 
 Much more powerful than analog synthesizers, both in terms of parameter 
 richness and control sophistication, are musical systems based on digital 
 computers. Here, under the direction of appropriate programs, a computer is 
 instructed to simulate a series of instrument building blocks, such as digital 
 oscillators, filters, adders, multipliers, attack generators, etc., from which 
 the musician assembles the instruments for his musical piece. The computer 
 calculates and outputs the desired sound waves in binary sampled-data form. 
 Hence, D/A converters must be used to convert the samples into electrical 
 pulses, and appropriate low-pass filters are needed to smooth the pulses and 
 produce the continuously varying electrical voltages suitable for driving a 
 system of loudspeakers. With the power of modern digital computers it is not 
 difficult to envision the synthesis of virtually any sound that could come 
 from a loudspeaker, which is a fairly general sound source available today. 
 Therefore, the sound qualities obtainable by the computer method span a much 
 richer range than in the case of analog synthesizers. Having at his disposal 
 the resources of a whole computer, the musician can also make use of much 
 
more sophisticated algorithms for the control of musical parameters and the 
 organization of compositional elements. 
 
 However, when it comes to real-time performance capabilities, computers, 
 
 like analog synthesizers, suffer from severe limitations, albeit of a dif- 
 
 7 
 ferent nature. Apart from time delays determined by the particular mode of 
 
 operation of the computing center — batch processing, off-line D/A conversion, 
 etc. — there are certain limitations which are inherent to the computer itself. 
 Since the computer is actually controlling the air-pressure that is coming 
 directly from the loudspeaker, it must reconstruct every minute portion of 
 the acoustic waveform of the final sound, a computational task that involves 
 not only a lot of resources, but also a lot of time. While with analog syn- 
 thesizers the generation of waveshapes in the audio range is a rather trivial 
 matter that presents no real-time problems, with computers this same task 
 becomes the most demanding in terms of resources and computation time. Thus, 
 even if the musician had at his disposal a dedicated, general -purpose computer 
 and was allowed to communicate with it in an interactive mode, there is very 
 little he could do in real-time, just because the computer is unable to keep 
 up with all the computations that need to be carried out on that time scale.* 
 
 If we could somehow relieve the computer from the burden of computing 
 the waveshape samples, and were to use it only to control the perceptual 
 characteristics of musical sounds like timbres, pitches, durations, loudness 
 
 crescendi, tempi, etc., then it would certainly be able to keep up with the 
 
 7 
 necessary computations in real-time. This is so because the rates at which 
 
 these psychological events occur or change are much lower than the rates at 
 
 which the air-pressure vibrates. 
 
 * For a more detailed description of the equipment and techniques avail- 
 able to the electronic musician today, see ref. 8. 
 
To conclude, it seems that digital systems are particularly suited to 
 musical parameter control in real-time, while their real-time capabilities 
 for sound synthesis are rather limited. Analog synthesizers, on the other 
 hand, while incorporating only rudimentary facilities for parameter control, 
 can cope very easily with the generation and modification of acoustic wave- 
 shapes in real-time, although the ranges in sound quality are not as rich as 
 in digital synthesis. If one could improve the sophistication of analog sound 
 generators, it would seem that a satisfactory musical system with real-time 
 performance capabilities can be arrived at by hybrid techniques, where analog 
 circuitry is used primarily for sound generation, and digital circuitry is 
 devoted to parameter control. This, indeed, is the idea at the basis of the 
 system to be described next. 
 
 1. 3 General Concept of New Instrument . 
 
 If the assurance of a real-time interaction between composer/performer 
 and instrument is to be the primary objective, two conditions must be met: 
 both operator and machine must be capable of carrying out their respective 
 tasks in real-time. 
 
 Real-time capabilities for the machine are achieved by resorting to a 
 hybrid configuration, as mentioned in the previous section. Thus, the genera- 
 tion of sounds and, in general, the performance of those tasks that, because 
 of the great amount of computation associated with them, take too long with 
 digital computers, are assigned to analog circuitry. The exercise of control, 
 on the other hand, is carried out more efficiently with digital circuitry. To 
 make parameter control possible, the characteristics of analog circuits are 
 externally programmable. The programming is done directly by the digital por- 
 tion of the system, under the control supervision of the composer /performer. 
 
Knowing the limitations of analog synthesis with respect to sound quality, 
 one may argue that a hybrid system, as far as this musical feature is concerned, 
 is going to be limited at least by the extent to which the analog portion is. 
 This is indeed true. However, with the help of state-of-the-art technology 
 and careful design, the sophistication of sound generating and modifying cir- 
 cuitry can be improved considerably. A constant effort in this direction has 
 been made throughout the development of the system, from the design of the 
 pitch generators to the design of the timbre control, loudness control, sound 
 distribution system, and others. Due to the presence of analog circuitry 
 another problem arises, namely, parameter stability. The human ear is very 
 sensitive to effects that result from subtle relationships and proportions 
 among certain musical quantities: tuning, beats, dissonances, simultaneity 
 of events, etc. These psychoacoustical factors make a severe demand upon 
 analog circuit performance, a problem usually not so serious in all -digital 
 systems. In order to alleviate these shortcomings as much as possible, par- 
 ticular emphasis has been placed on the maximization of accuracy and stability, 
 especially where the demand for these features is more critical. 
 
 Once the conditions for real-time operation of the machine are satisfied, 
 we must ensure that the human operator, too, be able to keep up with his 
 tasks on the improvisational time-scale. For this to be possible, he must be 
 relieved of the burden of too much control detail, which requires an amount of 
 activity that is usually incommensurate with human efficiency. Human opera- 
 tion in real-time can be best achieved if the relationship of the performer/ 
 composer to the system is like that of a conductor to his orchestra, rather 
 than that of a player to his instrument. Instead of taking care of every 
 elemental musical constituent, as one would normally do with electronic music 
 instruments and, to a lesser extent, with traditional instruments, the impro- 
 viser should mainly limit himself to the activity of steering, guiding, and 
 
e 
 
 influencing the evolution of a system that is already, so to speak, capable 
 of playing by itself. 
 
 If the performer is to expect such a degree of cooperation from the sys- 
 tem, the latter must be provided with the capability of controlling its own 
 parameters automatically. The dynamic control of compositional elements — be 
 it automatic or under the immediate responsibility of the operator — requires 
 the generation of a continuous stream of binary information. Information for 
 atuomatic control is generated within the system by means of binary sequence 
 generators based on feedback shift-registers and read-write memories. 
 
 Besides being capable of automatic control, the system must also be re- 
 ceptive to some form of external control from the operator. Since our goal 
 is that of relieving the latter from too much control detail, the compositional 
 elements that are best suited to the direct control of the improviser, both in 
 terms of their musical significance and the associated data rate, are those 
 affecting music at the macrostructural level. The control of microstructural 
 elements, such as individual fluctuations in the perceptual characteristics 
 of sounds, involves a data rate that is generally too high in relation to 
 human capabilities in real-time, and is therefore handled more efficiently by 
 the system itself. The operator, however, can still affect microstructural 
 elements indirectly through the usage of the control power he has at the macro- 
 structural level. 
 
 The above concepts can be better illustrated by examining the stream of 
 control information involved in the production of a typical musical piece. 
 As already mentioned, the dynamic control of perceptual characteristics of 
 sounds is handled with sequences of binary control words. Typically, per- 
 ceptual characteristics fluctuate at rates of the order of 1 to 100 times 
 per second, which clearly shows that the operator can hardly keep up with the 
 
9 
 
 associated data rate, especially if several perceptual parameters are to be 
 controlled simultaneously. Hence, the generation of this type of control 
 sequences is delegated to the system, which can certainly keep up with the 
 required rate without difficulties. 
 
 Examination of the stream of information over a longer period of time, 
 as, for instance, over the span of a complete musical gesture, is likely to 
 reveal the presence of another class of fluctuations which affect the infor- 
 mation at the level of entire sequences. Sequence rests and activations, 
 fluctuations of sequence lengths and rates, changes in the ranges spanned 
 by sequence data words are examples of this class of macroscopic, or second- 
 order fluctuations which affect music at a higher level of structural com- 
 plexity. The manipulation of macrostructural elements itself presupposes 
 the need for a corresponding class of control sequences to carry out the 
 task, thereby leading to the notion of sequences controlling other sequences. 
 Owing to the multilevel structural complexity of music, the chain of sequences 
 controlling sequences of the level below can be expanded further. However, 
 we need not elaborate on this topic here, as our goal is directed solely to 
 pointing out the real-time implications of such a distinction between dif- 
 ferent classes of sequences. In this respect, what characterizes macroscopic 
 fluctuations is the lower rate at which they take place, which may be typically 
 of the order of one event every 1 to 100 seconds. This is of the same order 
 of magnitude of the rate at which a human can comfortably perform control 
 actions in real-time. Therefore, it seems rather appropriate to give the 
 performer direct access to the control of this class of fluctuations. It may 
 be noted, in passing, that the notion of sequences controlling other sequences 
 allows a more uniform treatment of control and a higher degree of standardiza- 
 tion in circuit design and implementation. 
 
10 
 
 The generation of sequences for the control of macrostructural fluctua- 
 tions need not be a prerogative of the performer exclusively. Indeed, in 
 the course of an improvisation he may be willing to seek the automatic co- 
 operation of the system in carrying out even this type of task. To ensure 
 maximum flexibility, it is desirable that the balance between man and machine 
 be allowed to vary continuously and under the control of the operator himself. 
 This feature brings about a distinction between automatic and manual informa- 
 tion which is instrumental in the design concept of the system. The first 
 refers to the information produced within the system by sequence generators, 
 as already mentioned. The second refers to the information that the operator 
 feeds into the system through the input interface. Whether at any moment a 
 given compositional element is to be controlled automatically or manually is 
 decided by the performer with the aid of a simple information-steering device. 
 
 l.U System Organization . 
 
 Figure 1 shows an overall system block diagram. Both in analog and digi- 
 tal terms the instrument can be partitioned into four distinct layers or sub- 
 systems. These subsystems, however, are not totally independent of each 
 other, since they are allowed to share common resources, to exchange infor- 
 mation, and to be put in hierarchical relationships with respect to each 
 other. This system partition is intended to reflect the musical notion of 
 four distinct orchestras which can play together and interact in a variety 
 of ways and at different levels. As a result, the performer is given more 
 flexibility in achieving contrasting musical situations than with a mono- 
 lithic, homogeneous system. 
 
 A common resource that is always allocated to one orchestra at a time 
 is the control panel, which constitutes the interface between performer and 
 instrument. The panel consists of an array of touch-sensitive switches with 
 
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 memory, and a corresponding array of light indicators to display the switch 
 states. The array of touch-switches* acts as a buffer memory to handle the 
 digital information flowing between operator and instrument. The control 
 panel serves a twofold purpose: to give the operator a means for setting up 
 his control information in the proper digital format for input into the system, 
 and to allow the monitoring of certain information generated within the system, 
 via the light indicators. Thus, the states of the touch-switches can be con- 
 trolled in a mutually exclusive manner, either by the operator or by the sys- 
 tem, and the proper information routing is done by the operator. 
 
 Due to the large number of switches required to perform all the desired 
 control functions, it is more practical to have a control panel with just 
 the array of switches and lights required by one orchestra only, and to time- 
 share the usage of the array among all four orchestras. Hence, facilities are 
 provided for demultiplexing the information flowing between the panel and the 
 four subsystems, and for multiplexing the data traveling in the opposite 
 direction.** The time-sharing of the panel not only results in reduced cost 
 and size, but also introduces more clarity and order in the control activity 
 of the performer. 
 
 The basic timing of the various control functions is governed by four 
 digitally-controlled oscillators (or clocks), one for each orchestra. Dif- 
 ferent control functions, even if pertaining to the same orchestra, in general 
 require different timing scales and different timing configurations, depending 
 on their musical significance. Hence, each basic oscillator must be properly 
 scaled and processed in order to obtain all the desired subtiming signals. 
 
 * The design of the touch-switch, which is centered around a pair of 
 cross-coupled NAND gates with high-beta transistor input buffers, is due to 
 Dr. J. Divilbiss. 
 
 ** The Multiplexer/Demultiplexer was designed by T. Noggle and R. Borovec, 
 
13 
 
 The simplest mode of operation is the one in which all four "basic clocks 
 run independently of each other and, therefore, each orchestra plays accord- 
 ing to its own tempo. Provisions are made, however, for the slaving of clocks 
 to each other according to certain rules, so that the timing systems of dif- 
 ferent orchestras can be put in hierarchical relationships. A meaningful 
 slaving rule,* although not the only one, is that in which, with the clocks 
 numbered in ascending order, say from to 3, each clock can be slaved to 
 any one or combination of the lower -numbered ones, but not to any of the 
 higher-numbered ones, as shown in Figure 2. Thus, clock 2, for instance, 
 can run independently, or can be slaved to clock 1, or to clock 0, or to clocks 
 1 and 0, but not to clock 3. The programming of slaving patterns offers a 
 good example of manual vs. automatic information usage. Whether at any 
 moment a given clock is or is not to be slaved to one of the other eligible 
 clocks is decided either directly by the performer, or automatically by the 
 system. It is the performer, however, who selects the mode of operation, that 
 is, who decides whether the slaving is to be controlled manually or automat- 
 ically. This mechanism is readily implemented in terms of 2-to-l data selec- 
 tors. 
 
 Since the automatic control of the hierarchical slaving is a decisional 
 problem that encompasses the individual orchestras, the system is provided 
 with a supervisor control unit to carry out this and other tasks of similar 
 type. The basic timing for this unit itself is derived from a combination of 
 the four clocks already mentioned. 
 
 Each of the four orchestras consists of two general -purpose instruments, 
 or voices, as they are called in electronic music jargon. Two of the orches- 
 tras include also a percussion ensemble of l6 instruments each. 
 
 * Conceived by S. Martirano. 
 
Ill 
 
 Figure 2. Clock Hierarchy. 
 
15 
 
 A voice module is shown in block diagram form in Figure 3. It consists 
 of a Frequency Synthesis device for the manipulation of pitches; a dual 
 Digital Waveshape Generator for the programming of timbres ; a Mixer/Modulator 
 and a Programmable Filter for the dynamic control of tonal characteristics; 
 an Attenuator/Locator for the instrumental as well as spatial characteriza- 
 tion of sounds. 
 
 The eight voice modules are coupled with each other in a variety of 
 ways , so that , under command from the control system — be it manual or auto- 
 matic — the electronic signals associated with one voice device can be uti- 
 lized to modulate parameters of others. For instance, one waveshape may be 
 used to modulate the amplitude, the pitch, or the timbre of another. This 
 form of analog parameter control is a feature that the system offers in 
 addition to the digital form of control already mentioned. Indeed, since 
 the whole system is hybrid, it comes as no surprise that control can be 
 exercised in digital as well as in analog form whenever this is possible 
 and musically effective. 
 
 Although percussive sounds can be readily synthesized by properly pro- 
 gramming the parameters of voice modules, they are actually generated sepa- 
 rately by means of simpler, special-purpose circuitry. This solution is 
 intended to reduce the overload of the voice modules, so that they can be 
 more efficiently allocated to the synthesis of sounds of higher complexity. 
 
 Almost all circuits are dc coupled, a feature that allows a certain 
 amount of circuit standardization. Thus, the basic clocks are nothing else 
 but waveshape generators as in the voice modules, the only difference being 
 in the values of timing capacitors, which are properly scaled in order to 
 obtain the desired parameter ranges. In this case the Frequency Synthesis 
 device serves the purpose of controlling durations and associated timing 
 functions, while the sub-audio waveshapes here obtained are utilized to 
 
16 
 
 WAVESHAPE 
 GENERATOR 
 
 FREQUENCY 
 SYNTHESIS 
 
 PROGRAMMABLE 
 MIXER/MODULATOR 
 
 PROGRAMMABLE 
 FILTER 
 
 PROGRAMMABLE 
 ATTENUATOR/LOCATOR 
 
 WAVESHAPE 
 GENERATOR 
 
 t t I T 
 
 QUADRAPHONIC OUTPUTS 
 
 Figure 3- Voice Module Block Diagram. 
 
IT 
 
 control slow-changing musical effects and parameters like crescendi, glis- 
 sandi, transpositions, timbres, portamento, etc. (Figure k) . 
 
 Another resource shared by all four orchestras is the output system which 
 carries the sounds created within the instrument to the outside world. For 
 performances confined to a limited space or for studio recordings , the instru- 
 ment operates in a quadraphonic mode. Each orchestra controls the directional- 
 ity of its sounds dynamically by continuously programming their distribution 
 among the four channels. This task is carried out by means of programmable 
 joystick-like devices, which constitute part of the Attenuator/Locator appear- 
 ing in the block diagram of Figure 3. Facilities are also provided for the 
 control of the depth of sounds, besides their directionality, through the use 
 of programmable artificial reverberation. 
 
 For performances taking place in a concert hall, the spatial resolution 
 of the quadraphonic system can be made more subtle and sophisticated by resort- 
 ing to a larger number of loudspeakers properly distributed among the audience. 
 In this case, the output system is switched to an alternative mode of opera- 
 tion which involves the use of 2k loudspeakers having k input channels each — 
 one for each orchestra — for a total of 96 channels, as shown in Figure 5. 
 The presence or absence of sound on any of these channels is controlled by a 
 corresponding audio gate. Thus, every orchestra can control the routing of 
 its own sounds to any combination of the 2k loudspeakers according to patterns 
 that can be programmed directly by the performer on the control panel. Such 
 spatial effects as sound travel and spatial dialogues can be easily imple- 
 mented with this type of sound distribution. 
 
18 
 
 SEQUENCE 
 CLOCKING 
 
 PROGRAMMABLE 
 
 BINARY SEQUENCE 
 
 GENERATOR 
 
 \7 
 
 SUB-AUDIO 
 FREQUENCY 
 SYNTHESIS 
 
 SUB-AUDIO 
 
 WAVESHAPE 
 
 GENERATOR 
 
 DURATION 
 CLOCK 
 
 SUB-AUDIO 
 WAVESHAPE 
 
 Figure h. Duration Clock Generation for One Orchestra. 
 
19 
 
 OUTPUTS FROM 
 FOUR ORCHESTRAS 
 
 V V V V 
 
 AUDIO 
 GATES 
 
 LOUDSPEAKERS 
 
 vV\ 
 
 N 
 
 #1 
 
 I I I 
 
 I I I 
 
 I I I 
 
 </ 
 
 
 #24 
 
 Figure 5- 96-Channel Output System. 
 
20 
 
 II. TONE SYNTHESIS 
 
 2.1 Harmonic Tone Generation . 
 
 Harmonic tones are periodic sounds consisting of a mixture of sinusoidal 
 waves — also called pure tones — whose frequencies are integral multiples of a 
 basic frequency called the fundamental. Since these sounds play a rather 
 important role in music, it comes as no surprise that their generation and 
 control constitute the central task of the instrument under discussion. 
 
 The most important subjective qualities of harmonic sounds are pitch 
 and timbre. Pitch is the perceptual counterpart of the fundamental frequency, 
 to which it is related by a logarithmic dependence. Timbre is the result of 
 the distribution of the relative amplitudes and phases of all the significant 
 components of a tone, and is, therefore, the subjective counterpart of the 
 spectral structure of sounds. 
 
 Important in music are not only the tonal qualities a sound exhibits at 
 a given moment, but also the manner in which these qualities evolve with 
 time. Hence the versatility of a harmonic tone generator is directly re- 
 lated to the range of tonal possibilities it can achieve, together with the 
 amount of sophistication it offers for the dynamic control of the tones. 
 
 The usual method of generating harmonic tones is by means of relaxation 
 oscillators that incorporate some form of voltage control for the frequency 
 
 Q 
 
 of oscillation. The waveshapes generally available are the saw-tooth, tri- 
 angle, square-wave, pulse and sine. Musically there is no reason to prefer 
 these waveshapes to others. The choice of this particular set is dictated 
 by electronic convenience: once one of the first two waveforms is available — 
 and this is usually the case with relaxation oscillators — all others can be 
 easily obtained from it by means of relatively simple circuits. Additional 
 
21 
 
 waveforms can be generated by mixing waves of the basic set together, or by 
 altering their spectral structure through the use of formant filters or non- 
 linear signal processors. Although expanding the initial set of basic wave- 
 shapes, these techniques afford an amount of tonal variety and a sophistica- 
 tion of dynamic tonal control that are still inadequate to span a broad 
 musical scope. 
 
 In view of this limitation, an alternative approach to the synthesis of 
 harmonic tones was adopted here, namely, the generation of waves in sampled 
 form. This method, as shown in the block diagram of Figure 6, is centered 
 around a binary counter and a random-access memory. The waveshape is digi- 
 tized and stored in the memory, with each memory word holding the value of 
 one waveshape sample. Memory addressing is provided by the binary counter, 
 whose modulus equals the total number of words in memory. A train of pulses 
 feeding the counter results in a sequential, repetitive scan of all memory 
 words which are thereby displayed at the memory sense outputs and converted 
 into analog voltages by a D/A converter. Thus, what is ultimately obtained 
 is a staircase version of the waveshape stored in memory. In order to elimi- 
 nate the unwanted discontinuities introduced by the finite sampling resolu- 
 tion, the staircase is subsequently smoothed by a low-pass filter. 
 
 It is immediately seen that pitch can be controlled merely by program- 
 ming the frequency of the pulse train feeding the counter. To maintain a 
 uniform amount of smoothing that is independent of pitch, the position of 
 the filter cut-off frequency relative to the fundamental frequency of the 
 waveshape must be kept constant over the whole frequency range. Thus the 
 filter is also programmable, and its tuning is controlled by the same 
 mechanism that programs the frequency of count. 
 

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 The shape of the wave emerging from the filter is determined by the 
 information residing in memory. Being provided with read/write capabilities, 
 the memory can "be reprogrammed at will to accommodate new waveshapes. Hence, 
 the same basic circuitry lends itself rather easily to the generation of a 
 rich family of harmonic tones. 
 
 Usually a memory-write operation results in an abrupt change from one 
 waveshape to another. From the viewpoint of tonal control, however, it is 
 generally desirable that waveshapes undergo smooth transitions, although 
 abrupt changes may be occasionally acceptable as an option. 
 
 If the initial and final waveshapes are available simultaneously, a 
 smooth transition between the two can be easily implemented in analog fashion 
 by means of a continuously programmable mixer. The simultaneous generation 
 of two waveshapes is achieved by using two separate memories instead of one. 
 To ensure that the waves emerging from the corresponding D/A converters have 
 identical pitch, albeit different shapes, both memories are addressed by the 
 same counter. The two waveforms are then fed to the analog mixer where they 
 are combined in programmable, complementary percentages. Thus, the applica- 
 tion of a linear sweep to the mixer programming input causes the emerging 
 waveshape to undergo a smooth glide from one of the incoming waveshapes to 
 the other. At the beginning or at the end of a sweep, therefore, the contri- 
 bution to the mixer output comes entirely from one of the two memories, while 
 the contribution from the other is under complete attenuation by the mixer 
 itself. By allowing either memory to be updated only when its contribution 
 is totally attenuated, the discontinuities associated with memory-write 
 operations are always kept below audibility. Consequently, the listener 
 perceives waveshape changes only in the form of smooth transitions. 
 
2U 
 
 This technique of dynamic spectral control can be put to good use, 
 among others, for enhancing the expressiveness of synthesized sounds. It 
 has long been recognized that electronic sounds generally lack the warmth 
 and life that characterize natural sounds. To a large extent this dif- 
 ference stems from the fact that the spectral structure of a natural sound 
 
 Q 
 
 changes considerably in the course of its duration, however brief it may be. 
 
 While computer sound generation does not lack effective approaches to the 
 problem of spectral control, the techniques employed in analog synthesis 
 are somewhat artificial and unsatisfactory. It is seen that the hybrid 
 approach to sound synthesis here adopted allows for flexible control over 
 the dynamics of audio spectra. 
 
 The versatility of the waveshape generator can be further expanded by 
 allowing for alternative modes of memory read/write control. Thus, facili- 
 ties are provided for overriding the constraints illustrated above and for 
 updating a memory even when the corresponding output is not necessarily under 
 complete attenuation by the mixer. This option is particularly useful in 
 view of the fact that , when in the write-mode , the information present at the 
 data inputs of the memories here used is transferred directly to the sense 
 outputs. If the input information is derived, for instance, from a pseudo- 
 random binary generator, then one obtains colored noise, and the amount of 
 coloring can be easily controlled with the programmable, low-pass filter 
 available at the output. By forcing only one of the memories into the write 
 mode while the other is allowed to proceed in the usual way, the mixer output 
 consists of pitched noise, and the amount of pitching can be programmed to 
 arbitrary values with the aid of the mixer itself. 
 
 These options can be properly exploited for the creation of a variety 
 of special effects, as will be discussed in more detail below. Also below, 
 
25 
 
 it will be shown how the mixer, besides serving the purposes illustrated 
 above, can be used to implement various forms of amplitude modulation, there- 
 by expanding the class of available sounds to include anharmonic ones. 
 
 2.2 Pitch Control . 
 
 The purpose of the pitch control circuit, as shown in Figure 7, is the 
 generation of a voltage signal in the range to +10V, which will ultimately 
 be mapped into a pitch in the audio range. 
 
 Seven bits of information yield 128 different pitches which, in the 
 basic mode of operation, implement an equitempered scale of 12 tones per 
 octave. Besides this basic mode, the system can operate also with l6, 20, 
 or a variable number of equitempered tones per octave. The variable number 
 can be any integer or non-integer value between l6 and 20. All of these 
 options could be readily implemented by means of a multiplying D/A converter, 
 However, because of the still prohibitive prices of these devices at the 
 time of design — 1972 — a D/A converter of fixed full-scale was used instead, 
 and the proper scales are obtained with the help of a digitally -programmable 
 amplifier in conjunction with an external, variable-transconductance , analog 
 multiplier. Independently of the equitempered mode in which the system is 
 operating, the circuit also incorporates inputs for uniform pitch transposi- 
 tion and for frequency modulation. 
 
 The reference voltage to set the scale of the D/A converter as well as 
 the scales of the circuits to be described in the following two section are 
 derived from the temperature-compensated 10V voltage source internal to the 
 D/A converter itself. 
 
 2.3 Exponential Current Generation . 
 
 Owing to the fact that the frequency intervals of interest in music 
 form a geometric rather than an arithmetic progression, the linear domain 
 
26 
 
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 of the pitch control-voltage just discussed must be mapped into an exponen- 
 tial domain. This conversion is accomplished by exploiting the relationship 
 between collector current I_ and voltage drop v^„ across the base-emitter 
 
 junction of a silicon transitor, which is expressed by the well known for- 
 
 , 11,12 
 mula 
 
 I c = a F I Eg [exp (qv BE /kT) - l] . (l) 
 
 Here a„ is the forward, short-circuit current gain, common base; I is the 
 r lib 
 
 emitter saturation current; q the electron charge; k Boltzmann's constant; 
 T the absolute temperature. 
 
 The dynamic range of interest for pitches is about 3 decades, or 10 
 octaves. For reasons that will become apparent later, the corresponding 
 range for exponential current I has been chosen to lie between 1 uA and 
 1 mA. For a high-beta transistor operating over this range, Equation (l) 
 can be replaced, at the price of negligible error, by the truly exponential 
 relation 
 
 I c = I Eg exp (qv BE /kT) . (2) 
 
 Since the human ear is extremely sensitive to pitches out of tune, the 
 utmost care must be exercised to ensure a stable and accurate exponential 
 generation. Thus, compensation for temperature variations of I c is achieved 
 
 by resorting to a pair of transistors lying on the same substrate and, there- 
 
 12 
 fore, satisfying the conditions for close matching and thermal coupling. As 
 
 shown in Figure 8, the exponential conversion proper is done by transistor Q3. 
 Transistor QH , being driven at constant current by regulator Al , develops a 
 base-to-emitter voltage drop which is also constant, apart from temperature 
 variations tracking analogous variations in Q3. Thus QU provides the appro- 
 priate low-impedance, temperature-tracking emitter drive for exponential 
 generator Q3 . 
 
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 Compensation for temperature variations of I does not eliminate 
 thermal fluctuations entirely, however. As Equation (2) shows, I still 
 depends on T through the exponential term. To minimize this last source of 
 instability, the temperature of the transistors' substrate is maintained 
 constant by another regulator circuit. The RCA 30h6 transistor array, be- 
 sides the pair assigned to the exponential function generation, includes 
 three additional transistors. One of these, Q5 , is connected as a diode 
 and is used by external regulator A2 to sense the IC substrate temperature. 
 The remaining two, Ql and Q2 , merely dissipate power in order to keep the 
 substrate temperature constant at a value set by external trimpot R6. This 
 value is chosen so that the quiescent power dissipated by Ql and Q2 into the 
 substrate, at normal ambient temperature, is half its maximum. Transistors 
 Ql and Q2 draw about 30 mA of current and operate in a constant-current mode 
 in order to avoid large load changes for the positive power supply. 
 
 Having minimized thermal instability, there remains one last source of 
 
 error to be taken care of. This is introduced by the bulk resistance r„ of 
 
 hi 
 
 the emitter region of Q3 , which causes the junction voltage v to differ 
 
 Bhi 
 
 from the voltage V actually measured between base and emitter terminals, 
 by the amount r„I_. If this fact is taken into account, then Equation (2) 
 
 ill O 
 
 becomes 
 
 I c = I £S exp [q(V £E - r £ I c )/kT] , 
 
 which shows that the relationship between I and V is not truly exponential. 
 
 As r is only of the order of 10 ohms, the bulk voltage r I p is inconsequential 
 
 in the lower part of the I range. Not so in the upper portion of the range, 
 
 where, for instance, r_I_ is no longer negligible in comparison with a V^^ 
 
 ill L, kit, 
 
 excursion of about 18 mV, which is the amount typically required to effect a 
 change of one octave in I„. 
 
 v 
 
30 
 
 This source of inaccuracy can be compensated for by applying a correc- 
 tive voltage of magnitude r I either to the base of Q3, in the positive 
 direction, or to the base of Ql+ , in the negative direction. As a negative 
 voltage proportional to I is already being developed across emitter resis- 
 tor R3, the desired correction is achieved merely by scaling such voltage 
 down and by feeding it to the base of QU, a task accomplished by resistor 
 divider Rh and R5. 
 
 Trimmers Rl and R2 set the width and the base of the I range respec- 
 tively. Zener diode Dl clamps the upper value of I in the case of input 
 overdrive, and acts therefore as a filter for pitches beyond approximately 
 20 kHz. To avoid accuracy degradation, the exponential generator is buffered 
 to the subsequent stage by high-beta, common-base transistor Q6. 
 
 2.k Relaxation Oscillator . 
 
 The task of this stage is to convert the exponential current into a 
 proportional frequency which in turn is used to drive the counter address- 
 ing the memory. In order for the final waveshape to lie in the audio range, 
 the counter must count N times as fast as the intended audio frequency, where 
 N is the modulus of the counter. As the counter used is five bits long, 
 this means the exponential frequency must lie approximately between 6k0 Hz 
 and 6h0 kHz, a range which presents unusual problems if accuracy is to be our 
 primary objective. 
 
 Current-to-frequency conversion is achieved by means of the relaxation 
 oscillator shown in Figure 9« Assuming for a moment that Rl = and capaci- 
 tor C is discharged, current I., charges C and causes the voltage at the lower 
 side of C to decrease from the initial value of 6.2 V. This voltage is 
 buffered by the high input-impedance, unity-gain, FET amplifier to comparator 
 Al which compares it with a threshold voltage of 1.9 V as set by resistor 
 
31 
 
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32 
 
 divider R2 and R3. As the capacitor voltage hits the threshold, Al triggers 
 the one-shot which in turn causes switch Ql to close and discharge C. The 
 delay-constant of the one-shot is about 150 ns , which is long enough to allow 
 for the complete discharge of C by Ql. After the one-shot time-out, switch 
 Ql is opened again and the cycle is repeated. 
 
 The introduction of the one-shot between Al and Ql serves the purpose 
 of establishing a reliable delay between the triggering of Al and the 
 initiation of a new cycle. Thus, fluctuations of the time taken by Ql to 
 discharge C are completely masked by the stable delay-constant of the one- 
 shot. 
 
 The overall delay t occurring between the triggering of Al and the be- 
 ginning of a new cycle, although inconsequential at low frequencies, is no 
 longer negligible in the upper frequency range, where periods are of the 
 order of microseconds. This fact causes the relationship between output 
 frequency and input current I to be no longer linear. The error intro- 
 duced by the presence of x can be compensated for by causing comparator Al 
 to trigger a time t prematurely, so that the overall duration of a cycle 
 can be restored to its proper value. Such compensating action is performed 
 by resistor Rl. Being driven by current I , Rl develops voltage drop Rl'I- 
 in series with the voltage across capacitor C. If V is the capacitor voltage 
 that causes Al to trigger, then 
 
 6.2 V - V - Rl • I = 1.9 V . 
 
 The 
 
 time T it takes to charge C to voltage V at constant current I is 
 
 T c ■ c • \ n c ■ 
 
 Elimination of V yields 
 
 T Q = U.3 ' C/I c - Rl • C . 
 
33 
 
 The duration T of an entire cycle is then 
 
 T=T+x=U.3- C/I - Rl • C + t . 
 
 By choosing Rl so that Rl = t/C, the frequency f = l/T becomes 
 
 f = I C /(U.3 ' C), 
 i.e. , the frequency is linearly related to current I over the whole range. 
 
 It is worth noticing that, to a first-order approximation, the error 
 contributed to f by x is of the same type as the error contributed by the 
 exponential generator bulk-resistance r , as a series expansion of the cor- 
 responding formulas will quickly reveal. Thus, rather than having a separate 
 trimpot for the bulk-resistance compensation, a fixed resistor divider was 
 used there, and the final overall trimming for both compensations is done 
 by means of Rl alone. 
 
 In order to maintain the fluctuations of stray-capacitances negligible 
 in comparison with the value of capacitor C, the latter should be chosen as 
 large as possible. The larger the capacitor, however, the larger must be 
 current I , in order to obtain the prescribed frequency range, and this is 
 undesirable in view of the error caused by bulk-resistance r-g. A compromise 
 was achieved by choosing C = 330 pF and I in the range 1 yA to 1 mA. 
 
 To improve noise and jitter immunity, comparator Al drives the one-shot 
 through its Schmitt -trigger input, and Al itself is connected as a Schmitt- 
 trigger by virtue of diode Dl. 
 
 Transistor Q3 serves the purpose of interfacing the one-shot to switch 
 Ql, while transistor Q2 ensures proper start when power is turned on. 
 
 The overall accuracy of the oscillator frequency vs. pitch-voltage ex- 
 ponential relationship was measured to be better than + .2% over the pre- 
 scribed three decades of range. This figure is about l/30th of a semitone 
 in the conventional scale. 
 
3k 
 
 2.5 Digital Waveshape Generation . 
 
 Because of the discretization introduced by the sampling technique, the 
 control of tonal qualities can be exercised only over a limited bandwidth. 
 According to the well known sampling theorem, the order of the highest har- 
 monic that can still be represented with the sampling process equals half 
 the number of samples contained within one waveshape period. Thus, if the 
 harmonic generator is to span a wide tonal scope, the number of sampling 
 intervals should be as large as possible. A high sampling density, however, 
 requires a commensurably high frequency for the sequential scan of memory as 
 well as a large memory size to accommodate the samples. As it has already 
 been pointed out in connection with the relaxation oscillator, the accurate 
 generation of an exponential frequency over a range of three decades becomes 
 more and more difficult as the range is shifted in the direction of higher 
 frequencies. A compromise has been achieved with the choice of a time resolu- 
 tion of 32 samples per period and an amplitude resolution of l6 levels per 
 sample. As the tonal bandwidth is determined solely by the number of sam- 
 ples, the amplitude resolution need not be as high as the time resolution. 
 It should be noted, however, that the presence of the mixer results in an 
 effective increase of the amplitude resolution because it allows for the 
 programming of arbitrary ratios between corresponding samples of the incoming 
 waveshapes. 
 
 With the above specifications, the realization of a digital waveshape 
 generator requires four SNT^89 IC bipolar memories, as shown in Figure 10. 
 The open-collector memory outputs are fed to simple diode-resistor D/A con- 
 verters which generate the low-level analog signals — to approximately +32 un- 
 suitable for driving the programmable mixer. The inputs to the mixer are 
 provided in balanced form to eliminate the d.c. offset of about l6 mV 
 
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 introduced by the D/A converters and also to compensate for possible offsets 
 caused by mismatches between the input transistors of the mixer. 
 
 Figure 10 also shows the circuitry involved in the control of memory 
 read/write operations. The pair of LM 301 comparators with TTL-compatible 
 outputs constitutes a window detector having thresholds of IV and 9V re- 
 spectively. The detector continuously monitors the mixer programming vol- 
 tage w and allows either pair of memories to be updated only when the cor- 
 responding output is under complete attenuation by the mixer. Input v. is 
 attenuated when w is above 9V and input v when w is below IV. When w lies 
 inside the window, neither signal is completely attenuated and memory up- 
 dating is inhibited. This mode of operation is particularly desirable for 
 the production of continuous, smooth waveshape changes and is selected by 
 setting control bits CI and C2 both to logic 0. 
 
 When C1C2 = 01, memory-write operations are inhibited altogether, re- 
 gardless of the window detector response. This mode is useful for amplitude 
 modulation, as will be discussed in more detail in connection with the 
 mixer. Such an effect is implemented by programming the mixer with a modulat- 
 ing signal which causes the output waveshape to glide repetitively back and 
 forth between the two waveforms trapped in the respective memories. 
 
 When C1C2 = 10, the read/write mode of the upper memory pair is con- 
 trolled by the corresponding comparator in the usual way, while the other 
 comparator is overridden and the lower memory pair is permanently forced 
 into the write mode. This control state offers the option of bypassing one 
 of the memory pairs, so that the information present at the data inputs can 
 be injected directly into the mixer, via the D/A converter, where it is then 
 combined with the waveshape emerging from the other memory pair. This option 
 is made possible by the fact that the SNT^89 memories, when the write mode 
 
37 
 
 is enabled, display at the sense outputs the same information — albeit in 
 complemented form — that appears at the data inputs. 
 
 The usefulness of this control state is typified by the generation of 
 pitched noise, which is easily achieved by driving the memory data inputs 
 with a pseudo -random, binary number generator and by setting C1C2 = 10. 
 The presence of the mixer allows for the programming of arbitrary amounts 
 of pitching, while the output low-pass filter can be exploited to impart 
 different types of coloration to the sound emerging from the mixer. The 
 class of sounds thus obtained is particularly useful in the synthesis of 
 percussive sounds of the crash-cymbals or snare-drums type, or in the crea- 
 tion of special effects like steam gushings, surf surges, etc. 
 
 The control state C1C2 = 11 puts both memory pairs into the write mode 
 and causes them to be loaded with the same information. This mode is useful 
 for amplitude modulation with carrier suppression, as will be discussed in 
 more detail in connection with the mixer. 
 
 The information to be loaded into the memories during write-operations 
 is normally derived from a central, pseudo-random, binary number generator. 
 Since the amount of hardware and control is kept to a bare minimum, this form 
 of information generation offers the significant advantages of simplicity and 
 economy. It also offers a more unified approach to sound synthesis in that 
 the same circuitry can be used for the generation of harmonic as well as 
 noise-like sounds. Although this method, in conjunction with the control 
 options illustrated above, has proven quite adequate to span a vast range of 
 improvisational situations, there are circumstances which ask for a strictly 
 deterministic approach to the generation of waveshape information. To handle 
 these situations, a central memory bank is provided, where the information 
 pertaining to the waveshapes of interest is stored and made available for 
 
38 
 
 consumption by any of the harmonic generators. Thus, additional steering and 
 control facilities are provided for bypassing the binary noise-generator and 
 for properly routing the desired information. 
 
 A typical example of this class of situations is offered by the synthesis 
 of speech-like sounds. Being harmonic to a considerable degree, human vowels 
 are particularly suited to the digitized synthesis technique here described. 
 Since each vowel has a predetermined waveshape, the associated digital infor- 
 mation is stored in a central read-only memory from where it can be loaded, 
 upon request, into any of the peripheral waveshape generator memories. It is 
 worth pointing out, in passing, that the presence of the programmable mixer 
 offers a rather simple means for the generation of diphthongs, and that the 
 control options associated with pitched and colored noise can be put to good 
 use for the synthesis of a number of fricative consonants. 
 
 2.6 Programmable Mixer/Modulator . 
 
 The function of this circuit is to mix two input signals v and v in 
 
 A is 
 
 complementary percentages , with the mixing ratio being programmable by an 
 external control voltage w. Thus, if v_ denotes the mixer output, the de- 
 sired transfer function is 
 
 v Q = (w/10 V) v A + (1 - w/10 V) v B , ' 
 where control-voltage w spans the range to +10 V. 
 
 As shown in Figure 11, the mixer is centered around two variable- 
 trans conductance transistor pairs, very much like present day four-quadrant 
 
 13 lU 
 IC multipliers. ' However, unlike in a multiplier, the inputs to the 
 
 transistor pairs are kept separate and their collectors are connected in 
 phase. The bias current for powering the transconductance elements is pro- 
 vided by current generator Q8. After going through differential pair Q3/Q6, 
 this current is split into two complementary components which bias 
 
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 transconductance pairs Q1/Q2 and Q^/Q5 independently. As the gain of a 
 transconductance element is proportional to its bias current, the relative 
 contributions of the two pairs to the common output are determined by the 
 ratio of the respective bias currents. Hence, the mixing ratio can be 
 easily controlled by programming current splitter Q3/Q6. 
 
 The circuit parameters have been chosen so that a control -volt age span 
 of 10 volts slightly overdrives the current splitter. This precaution en- 
 sures that at the edges of the control range one of the transistors of the 
 current splitter is completely cut off and the corresponding transconductance 
 element brings no contribution to the mixer output. The transitions from 
 linear to saturating regions occur around 1 V and 9 V respectively, and take 
 place smoothly by virtue of the knee exhibited by the base-emitter junction 
 characteristics of Q3 and Q6. 
 
 The nature of the input stage of the programmable low-pass filter, to be 
 discussed below, allows for the balanced output lines of the mixer to be fed 
 directly to the filter with no need for additional interface circuitry. 
 
 Owing to the inherent multiplicative capabilities of the mixer, the 
 areas of application of this device can be further expanded to include vari- 
 ous forms of amplitude modulation. To this end, in addition to the input for 
 control voltage w, another terminal is provided for applying the modulating 
 signal. Both inputs are then added together by common-base transistor Q7- 
 
 When the circuit is operated as a modulator, waveshape updating is 
 usually inhibited (C1C2 = 01), owing to the fact that the purpose of the win- 
 dow detector is defeated by the presence of the modulating signal. The type 
 of modulation most commonly obtained is without carrier suppression, since 
 the circuit is basically a two-quadrant multiplier. However, the carrier 
 can be suppressed by having v. = -v_. and by d.c. offsetting the modulating 
 signal by 5 V, as can be readily verified from the mixer transfer function. 
 
Ill 
 
 These conditions are achieved by first setting control state C1C2 = 11, which 
 causes "both memory pairs to he loaded with the same waveshape. Subsequently, 
 control bit CI is changed to zero, while w is being set to +5 V. The required 
 phase inversion of one of the waveshapes is ensured by the fact that the 
 balanced outputs from the corresponding D/A converter are wired interchanged 
 to the mixer. This asymmetry does not disturb the other control modes since 
 the phases with which waveshapes are loaded into memories are immaterial. 
 
h2 
 
 III. ELECTRONICALLY PROGRAMMABLE FILTERS 
 
 A filter is electronically programmable if its characteristics — corner 
 or center frequency, damping factor, etc. — can be altered by means of external 
 control signals. Although a considerable amount of work in the area of active 
 filters of fixed or manually programmable characteristics has been going on for 
 more than a decade now, electronically tunable filters have made their 
 appearance only in very recent years. Filter parameter control is achieved 
 by means of devices exhibiting some form of multiplicative capabilities. 
 
 JFET voltage-controlled resistors, analog multipliers, ' and periodic- 
 
 19 20 
 switched filter networks with variable duty-cycle ' are examples of the 
 
 multiplicative components and techniques being presently used. All of these 
 
 methods, however, seem to suffer from one kind of limitation or another which 
 
 restrict their useful ranges of operation to no more than a few octaves. 
 
 The need for a filter of much wider programming range has led to the 
 development of two building blocks which, when properly utilized in the syn- 
 thesis of filters, allow for more than three decades of parameter control. 
 A range of this width is made possible by the fact that the controlling sig- 
 nals are currents, rather than voltages. Indeed, currents can be used to 
 accurately represent analog variables over much wider ranges than voltages, 
 a feature of present-day technology which never seems to be stressed enough. 
 In fact, by exploiting the highly predictable relationship between collector 
 current and base-to-emitter voltage drop of the silicon transistor, currents 
 can be easily controlled over a number of decades, while ordinary voltage 
 ranges are dynamically limited by temperature drift and other forms of noise. 
 
 The heart of the filter blocks in question is a programmable transcon- 
 ductance element, also referred to as an operational transconductance 
 
U3 
 
 amplifier, or OTA for short. The device resembles an ordinary operational 
 amplifier in that is also has one output and two input ports, the polarity 
 of the output signal being the same as that of the voltage at one of the in- 
 put terminals, and opposite to that of the other. However, unlike opera- 
 tional amplifiers, the OTA's output signal is a current, rather than a voltage, 
 and the transfer characteristic — which has therefore the dimensions of a 
 conductance — can be controlled externally by means of a current signal, also 
 referred to as a bias current I . . Thus, in addition to the three signal 
 
 -DXAo 
 
 ports mentioned above, the OTA has a fourth control port for the application 
 of Lm-.Q. The definition of the ideal OTA is illustrated in Figure 12. 
 
 The filter building block to be described first is provided by the cir- 
 cuit of Figure 13. Using Laplace transforms, 
 
 V q = I Q /(sC) = g m (V + - V")/(sC) 
 or 
 
 V = (V + - V")/st 
 o 
 
 where 
 
 t = C /gm = (0/K)/I BIAS . 
 
 Thus the transfer function of the circuit is that of a bipolar integrator, 
 with the constant of integration being externally programmable by means of 
 
 I T . . This simple building block can be readily incorporated into filter 
 
 21 22 
 
 configurations based on integrators, like the state -variable or the biquad 
 
 ♦ topologies. Then, by controlling I with an exponential current generator, 
 
 r>U\.o 
 
 the constant of integration, i.e. the natural frequency of the filter, can be 
 readily varied over a wide dynamic range. 
 
 As an example, Figure ik shows how two such integrators can be connected 
 together to synthesize a filter which exhibits simultaneously a low-pass and 
 a band-pass transfer function of constant bandwidth. By working with Laplace 
 transforms in the manner indicated above, one can readily verify that 
 
kk 
 
 BIAS 
 
 z. = z. = z 
 
 i i o 
 
 ^ = g m (V+ " V_) 
 
 g m = KI BIAS ' K = constant 
 [gj = n _1 , [K] = V" 1 . 
 
 Figure 12. Definition of Operational Trans conductance Amplifier (OTA) 
 
 oV, 
 
 ro 
 
 BIAS 
 
 Figure 13. Programmable Bipolar Integrator. 
 
h5 
 
 BIAS 
 
 BIAS 
 
 Figure ik . Two-integrator Programmable Filter 
 
 BIAS 
 
 Figure 15. Programmable Conductance , 
 
k6 
 
 and 
 
 where 
 
 V B.P. /V i = Sw o /(s2 + SW o /Q + W o 2) 
 
 V T /V. = l/(s 2 + sa> /Q + co 2 ) , 
 
 Li. x • 1 O O 
 
 "6 " 1/T " KI BIAS /C 
 
 is the natural frequency of the filter, and 
 
 a /Q = 1/(CR) 
 o 
 
 is the constant bandwidth of the band-pass function. Resistor R has heen 
 
 added to insert a loss and therefore ensure a non-zero damping factor. If 
 
 R is omitted altogether, the circuit oscillates and, as such, it can be used 
 
 as a programmable sine-wave oscillator with quadrature outputs. Conversely, 
 
 the replacement of R with a programmable conductance, to be discussed below, 
 
 offers the option of independent control over the Q or over the bandwidth of 
 
 the filter. 
 
 The other building block is offered by the feedback configuration of 
 
 Figure 15. The behavior of this circuit can be concisely described by means 
 
 of its y-parameters , which are 
 
 y ll = y 12 = °' y 21 = - g m' y 22 = +g m ' 
 However, its usefulness as a building block for the synthesis of programmable 
 filters can be better visualized by considering its equivalent circuit, which 
 is shown in Figure l6. The circuit consists of a current-controlled conduc- 
 tance in series with a voltage follower to account for the infinite impedance 
 as seen into port 1 of the original circuit. If two devices like the one of 
 Figure 15 are connected in parallel but with the ports interchanged, the re- 
 sulting circuit behaves like a programmable, floating conductance. Hence, it 
 is a simple matter to replace some or all of the resistors of a given filter 
 with programmable conductances and thus make the filter itself programmable. 
 
hi 
 
 BIAS 
 
 Figure l6. Equivalent Circuit for Figure 10. 
 
 ViO 
 
 BIAS 
 
 BIAS 
 
 BIAS 
 
 Figure 17. Three-pole , maximally- flat , low-pass programmable filter, 
 
U8 
 
 In most cases of interest, however, a single OTA per resistor is suf- 
 ficient, without the need for the parallel configuration just discussed, 
 which requires two OTA's instead of one. Indeed, the high input-impedance 
 presented by port 1, far from being a limitation or a nuisance, is often a 
 
 desirable feature since it facilitates network coupling, and it can also be 
 
 23 
 exploited to improve filter sensitivity. 
 
 Figure 17 shows one way of realizing a three-pole, maximally-flat, low- 
 pass programmable filter with three conductance building blocks of the type 
 just described. The filter uses also a unity-gain voltage follower to de- 
 couple the output of the third OTA from the feedback capacitor and outside 
 loads . 
 
 Figure 18 shows a circuit realization of the operational transconduc- 
 tance concept which could readily be put in integrated form since it involves 
 circuit elements of well established feasibility in commercial linear IC's. 
 The input stage, as in present-day analog multipliers, consists of a 
 balanced voltage-to-current converter (Ql through QU ) and a diode compressor 
 (Q5/Q6) to provide the appropriate base-drive for trans conductance pair 
 Q7/Q8. This configuration results in a substantial improvement for the 
 
 linearity of the transconductance pair, while drift and other noise factors 
 
 2k 
 are also minimized. Because of its Darlington-connected, differential am- 
 plifier configuration, the voltage-to-current converter presents a high input- 
 impedance which allows for an OTA to drive other OTA's directly, as shown in 
 Figures ih and IT. 
 
 The balanced collector currents of transconductance pair QT/Q8 are con- 
 verted into a single-ended difference output current by means of current- 
 mirrors Q9/Q10/Q11, 012/013/01^, and Q15/Q16/Q17- To achieve high output- 
 
 25 
 impedance, these current-mirrors make use of the Wilson configuration. 
 
k9 
 
 u 
 
 a) 
 
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 CH 
 •H 
 H 
 
 I 
 
 0) 
 
 o 
 
 -p 
 
 o 
 
 o 
 
 o 
 
 EH 
 
 O 
 
 •H 
 
 13 
 
 dJ 
 
 ft 
 o 
 
 CO 
 
 •H 
 
50 
 
 The value of K is established by the circuit parameters of the voltage-to- 
 current converter — bias currents and amount of emitter degeneration. A con- 
 venient parameter choice which conforms with present analog standards is the 
 one which results in K = 1/(10 V). 
 
 Although the complete circuit of Figure 18 is not available in inte- 
 grated form, an IC OTA of similar concept but without the input converter 
 and compressor is presently manufactured by RCA (CA3080). For this OTA im- 
 plementation, K = q/2kT, so that, at room temperature, K - 19.2 • 10 V 
 
 The CA3080 also requires that I flow into the current control terminal 
 
 £S-LAb 
 
 which is kept at a negative potential. 
 
 The smoother for the harmonic tone generator has been realized with two 
 CA3080 OTA's in the manner illustrated in Figure 19- This circuit realizes a 
 two-pole, maximally-flat, low-pass programmable filter. Due to the absence 
 of the voltage-to-current converter and diode compressor, signal conditioning 
 at the OTA's inputs is achieved by means of resistor attenuators, as shown. 
 The resistor values have been chosen so that the balanced outputs from the 
 programmable mixer can be tied to the inputs of the first OTA directly, with- 
 out the need for additional ancillary circuitry. Also, because of the finite 
 impedance presented by the attenuators, the output from each OTA must be 
 buffered by a high input -impedance decoupler. This task is performed by the 
 pair of FET-transistor source followers. The voltage-offsets associated 
 with the followers do not affect the dc level of the filter because each fol- 
 lower is placed inside the feedback loop of the corresponding OTA. 
 
 Also shown in Figure 19 is a dual exponential current generator which 
 provides the appropriate bias for both OTA's. The generator consists of 
 three matched pnp transistors from the CA308U array and a discrete, high- 
 beta npn transistor. Being forced to conduct a constant current — about 1 mA — 
 
51 
 
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52 
 
 by regulator QU , transistor Ql develops a base-to-emitter voltage drop which, 
 apart from temperature fluctuations, is also constant. Thus the emitter ter- 
 minal of Ql constitutes a low-impedance point and presents a fixed voltage- 
 offset with respect to the base terminal. This is just what is needed to 
 provide the proper emitter drive for exponential current generators Q2 and 
 Q3« Since the emitters of Ql , Q2, and Q3 are tied together, any voltage 
 excursion applied to the base of Ql is transmitted to the emitters of Q2 and 
 Q3 unaltered, by virtue of the regulating action of Qi|. Furthermore, tem- 
 perature variations of the base-to-emitter voltage drop of Ql tend to cancel 
 out analogous variations affecting Q2 and Q3 , thereby providing temperature 
 stabilization for the exponential generators. 
 
 Trimpot R2 sets the base and trimpot Rl the width of the exponential range, 
 One of the exponential generator inputs is fed with the Pitch Voltage control 
 signal, so that the position of the filter corner-frequency, relative to the 
 sampling frequency, is kept constant over the whole range of interest. The 
 other control input is used to perturb or alter this fixed constraint , there- 
 by providing an additional form of spectral control besides the techniques 
 previously discussed. 
 
53 
 
 IV. SOUND INTENSITY AND LOCATION 
 
 U.l Intensity and Location Control . 
 
 The circuitry described so far has dealt primarily with the synthesis 
 and control of pitches and timbres. Another parameter of considerable 
 psychoacoustical consequence is intensity. As in the case of tonal color, 
 particularly important in music are the dynamic characteristics of this para- 
 meter, also referred to as the sound envelope. These characteristics are 
 related to the manner in which sounds "build up and decay in time, and con- 
 stitute a significant clue toward the instrumental characterization of sounds. 
 Given the importance of this parameter, it is desirable that intensity con- 
 trol allow for a broad choice of sound envelopes in order to ensure a wide 
 spectrum of instrumental capabilities . 
 
 In analog systems , the dynamic control of sound intensity is realized 
 by means of programmable attenuators operating under the control of envelope 
 generators. The types of envelope functions usually available are triangular, 
 trapezoidal, or trapezoidal -with-overshoot , and such envelope parameters as 
 attacks, steady-states, and decays must be individually preset by hand. 
 These techniques, far from satisfying a broad range of instrumental situations, 
 compound real time performance in that they necessitate manual control. By 
 using hybrid techniques, as it will be shown below, both of these problems 
 can be approached in a more organic fashion that allows the control of com- 
 plex and minute deviations in a manner that parallels the domain of natural 
 sounds. 
 
 Another parameter that has become rather significant in the music of the 
 last two decades is the location and movement of sound within a musical envi- 
 ronment. For studio-type operations, the output of the system under discussion 
 
5U 
 
 consists of four discrete channels. The control of spatial parameters is 
 accomplished by dynamically programming the distribution of sounds among 
 the four channels, and by adding programmable amounts of artificial rever- 
 beration to the original sounds. 
 
 In hardware terms, sound enveloping and quadraphonic sound location are 
 handled uniformly by means of a programmable attenuator/locator circuit. 
 This device acts simultaneously as programmable attenuator and programmable 
 joy-stick. 
 
 k .2 Programmable Attenuator/ Locator . 
 
 Since the harmonic tone generator produces sounds of constant amplitude, 
 signal level control is realized independently by feeding the generator out- 
 put into a programmable attenuator/locator and by controlling the amount of 
 attenuation with an envelope generator. A block diagram of the attenuator/ 
 locator is shown in Figure 20. 
 
 Owing to the fact that the human ear responds to the amplitude of sounds 
 in a nonlinear manner, it is necessary that the linear domain of the envelope 
 voltage first undergo an appropriate nonlinear mapping. A mapping which is 
 musically adequate as well as electronically convenient is the exponential 
 one, and the related circuitry is shown in Figure 21. The exponential con- 
 version proper is performed by transistor Ql while transistor Q3 and regula- 
 tor Q2 serve the purpose of providing the appropriate emitter drive for Ql. 
 Thus the circuit is of the type already encountered, except for a certain 
 degree of simplification justified by the less stringent requirements of 
 level control. Circuit parameters have been chosen so that a to 10 V con- 
 trol input range is mapped into an exponential current range of about 50 db. 
 Although the circuit can easily afford a much wider range, 50 db constitutes 
 a convenient compromise that ensures adequate musical range without imposing 
 
55 
 
 
 
 
 FRONT/LEFT 
 t 
 
 
 FRONT/ RIGHT 
 A 
 
 
 BACK/LEFT 
 A 
 
 
 BACK/RIGHT 
 A 
 
 
 
 TRANSCONO. 
 MULTIPLIER 
 
 
 
 TRANSCONO. 
 MULTIPLIER 
 
 
 
 TRANSCOND. 
 MULTIPLIER 
 
 
 
 TRANSCOND 
 MULTIPLIER 
 
 
 
 
 
 
 
 
 
 SIGNAL 
 
 
 t 
 
 \ 
 
 
 L 
 
 A 
 
 
 
 A 
 
 
 j 
 
 ^ 
 
 INPUT 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 CURRENT 
 SPLITTER 
 
 
 
 CURRENT 
 SPLITTER 
 
 
 
 
 
 
 
 
 LEFT/RIGHT 
 
 
 i 
 
 V 
 
 
 
 - 
 
 1 
 
 
 
 CONTROL 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 
 FRONT/BACK 
 
 
 CURRENT 
 SPLITTER 
 
 
 
 CONTROL 
 
 ^ 
 
 
 
 
 
 > 
 
 < 
 
 
 
 ENVELOPE 
 
 
 EXPONENTIAL 
 CURRENT 
 GEN. TR 
 
 
 
 VOLTAGE 
 
 
 
 
 
 
 
 < 
 
 t 
 
 
 ) 
 
 I 
 
 
 
 
 
 SOUELCH 
 
 
 
 
 
 C 
 
 :oni 
 
 ROL 
 
 
 
 
 
 
 
 
 Figure 20. Block Diagram for Programmable Attenuator /Locator. 
 
56 
 
 EXPONENTIAL 
 
 CURRENT 
 ( 3/aA TO 1mA) 
 
 A 
 
 + 15V 
 
 ENVELOPE lOK 
 
 VOLTAGE > f— WW 
 
 OTO + 10V) 
 
 100X1 
 
 FROM 
 < POINT "R" 
 (-3.3V) 
 
 Ql 8 Q3 = 
 (2/5) X CA3046 
 
 Figure 21. Exponential Current Generator and Squelch Control. 
 
57 
 
 too stringent demands on envelope control. However, 50 db below full sig- 
 nal level does not offer enough attenuation when silence is desired. This 
 inconvenience is eliminated by means of a squelching circuit which senses 
 the input control voltage and completely shuts off the exponential generator 
 as soon as the input goes below a few tens of millivolts. This artifice 
 greatly expands the total effective width of the range, while still keeping 
 the useful control portion at about 50 db. 
 
 The exponential current thus obtained is subsequently split into four 
 separate components which are used to bias four corresponding variable- 
 transconductance transistor pairs, one for each output channel. The signal 
 input terminals to all four pairs are tied together. Since the gain of a 
 trans conductance pair depends on the amount of emitter bias current, it is 
 immediately seen that the output signal level from one of the pairs, rela- 
 tive to the levels of the others, depends on the percentage of exponential 
 current diverted to that pair. Furthermore, all four gains depend on a com- 
 mon scale factor proportional to the magnitude of the exponential current. 
 Thus the exponential generator determines the overall sound intensity and 
 the current splitters determine the relative intensity distribution among the 
 four channels. 
 
 The circuit schematic for the transconductance pairs and current split- 
 ters is shown in Figure 22. As in the case of the programmable mixer, the 
 choice of circuit parameters is such that the current splitters are slightly 
 overdriven by a control voltage of 10 V range. This precaution ensures that 
 at each end of the control range one of the current splitting transistors is 
 totally shut off, thereby providing full signal attenuation at the transcon- 
 ductance pairs biased by that transistor. This feature allows complete 
 channel isolation. 
 
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 Diode-connected transistor pairs Q19/Q20 and Q21/Q22 constitute a non- 
 linear voltage compressor whose purpose is the linearization of the current 
 splitters' transfer characteristics, a topic already discussed in connection 
 with OTA's. The proper bias voltages for the current splitters and the ex- 
 ponential current generator are provided "by the string of diodes shown at 
 the right of the circuit diagram. 
 
 To illustrate the above arguments with an actual example, consider 
 transistor pair Q17/Q18, which constitutes the front /back current splitter. 
 When the front /back control voltage is around OV, Q15 is cut off, so that 
 Q21 and Ql8 are also off. Thus all the exponential current is diverted to 
 the "front" transconductance pairs, while the "back" pairs are completely 
 cut off. If the control voltage is now increased, conduction is gradually 
 transferred from Ql6 to Q15. This is accompanied by a simultaneous increase 
 in the conduction of Q21 and Ql8 and decrease in the conduction of Q22 and 
 Q17. Thus, the exponential current is gradually diverted from Q17 to Ql8, 
 causing a gradual attenuation at the "left" channels and gradual amplifica- 
 tion at the "right" channels. Upon reaching 10 V of control voltage, Ql6 , 
 Q22 and Q17 are completely cut off, and the situation is now reversed, as 
 compared with the case in which the control voltage is around 0V. Similar 
 arguments hold for the left/right current splitters. 
 
 It is a known fact that transconductance pairs, particularly when im- 
 plemented with general purpose IC transistors, generate an undesirable 
 amount of noise which limits their usage in high-quality audio applications. 
 Among others, the noise level depends on the amount of emitter bias, and 
 usually increases with the latter. This effect, however, is not so critical 
 in the present application, owing to the fact that the attenuator/locator is 
 being permanently fed with a full-level signal from the tone generator. 
 
60 
 
 Thus, an increase in emitter bias increases not only the noise level, but the 
 signal level as well, thereby ensuring a satisfactory signal-to-noise ratio 
 over the entire dynamic range of interest. Subjectively, this ratio is 
 further improved by the well known masking effect of human hearing, where- 
 upon weak sounds — noise in this case — are partially masked by loud sounds 
 in the same frequency band. 
 
 The circuit realization of the entire attenuator/locator, inclusive of 
 the exponential generator, requires four IC transistor arrays, besides the 
 ancillary circuit shown. Two CA305^ IC's are needed to implement the four 
 transconductance multipliers and the left/right current splitters. One 
 CA30U6 implements the remaining splitter and the exponential generator, and 
 one more CA30^6 is needed for the diode compressors. Each transconductance 
 transistor pair is provided with a trimmer for input offset compensation. 
 
 To render the attenuator/locator suitable for driving standard audio 
 equipment, the balanced outputs from each transconductance pair must be con- 
 verted into a single-ended voltage signal. This is readily accomplished by 
 means of a differential operational amplifier, in the conventional manner of 
 analog multipliers. However, since all eight voice modules share the same 
 four output channels, it is not necessary to use an operational amplifier 
 for each of the 32 transconductance pairs. Rather, the outputs from all 
 eight voice modules are connected in parallel, with corresponding collectors 
 of corresponding transconductance pairs being tied together. The desired 
 double to single-ended conversion is then accomplished with four differential 
 amplifiers only. This circuit configuration results in the saving of 28 
 operational amplifiers. 
 
 Before being fed to output audio equipment, each channel is sent through 
 a spring reverberator and the spring output is mixed in programmable amounts 
 
6i 
 
 with the unreverberated signal. The associated hardware, involving standard 
 audio circuitry, will not "be described here any further. 
 
 1*.3 Envelope Generation . 
 
 To control intensity dynamically, the exponential generator discussed 
 above must be programmed with control functions of time, or envelopes. This 
 task is handled in hybrid fashion by means of an envelope generator in con- 
 junction with a binary word sequence generator. 
 
 Envelopes are generated in the form of joined straight line segments. 
 Thus the characteristics of a given envelope are entirely defined once the 
 locations of its breakpoints are known. The specification of a breakpoint 
 involves eight bits of information. Four bits of amplitude information 
 determine the breakpoint ordinate, and four bits of rate information deter- 
 mine the slope of the line segment terminating at that point. 
 
 The typical generation of a segment occurs as follows. The envelope 
 generator sends the binary word sequence generator a request for an 8-bit 
 word of data. After receiving the data, the envelope circuit generates a 
 voltage ramp at the rate specified by the homonymous bits. The polarity of 
 the ramp is implied by the remaining four bits of amplitude. Depending on 
 whether these bits specify a greater or a smaller voltage than that cur- 
 rently output by the envelope generator, the ramp polarity is respectively 
 positive or negative. Thus, in the course of ramp generation, the output 
 voltage approaches linearly the value specified by the amplitude bits. Upon 
 reaching this value, the generation of the current line segment is completed, 
 and a new data request is issued for the generation of another segment. 
 
 Four bits of amplitude as well as rate resolution may seem inadequate 
 to handle detailed envelope characteristics. It should be noted, however, 
 
62 
 
 that l6 loudness levels over a dynamic range of 50 db yield a resolution of 
 about 3 db , which compares favorably enough with the sound loudness resolu- 
 tion of the human ear. If a fine control over the envelope structure is 
 desired, this can be simply achieved by expressing the envelope function in 
 terms of small line segments. 
 
 The envelope generation process is illustrated in the block diagram of 
 Figure 23. The heart of the circuit is an operational amplifier with pro- 
 grammable slew-rate. This device is readily simulated with the help of 
 two ordinary amplifiers: a comparator to simulate the input stage, and a 
 current integrator to simulate the Miller stage. Slew-rate control is 
 achieved by programming the current feeding the integrator. To ensure a 
 wide dynamic range of envelope characteristics, the current is generated by 
 means of an exponential converter, as in the control of pitch and loudness. 
 
 When the output from the amplitude D/A converter exceeds the output 
 from the integrator, the comparator sets the current switch to the "0" 
 position. The integrator is therefore driven from the exponential generator 
 directly and produces a positive ramp, or attack. When the integrator output 
 exceeds the converter output, the comparator sets the switch to the "l" 
 position. Being now diverted to the current mirror, the exponential current 
 undergoes a polarity reversal, thereby causing the integrator to produce a 
 negative ramp, or decay. Thus, the chain comparator-switch-integrator con- 
 stitutes a voltage follower with externally programmable slew-rate. The 
 closed-loop frequency parameters of the chain are such that, when the output 
 becomes equal to the input, the comparator begins to oscillate between the 
 "0" and "l" states. This feature is intentionally exploited to mark the 
 achievement of an envelope breakpoint and therefore issue a request for a 
 new word of data from the binary sequence generator. The request generation 
 is handled by the breakpoint detector circuit. 
 
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 The detailed circuit diagram of the envelope generator, exclusive of 
 the binary sequence generator, is shown in Figure 2h . Binary data are 
 separately converted into analog signals by means of diode-resistor networks 
 and transistor switches, as shown. The rate D/A converter is fed to the 
 exponential generator, which consists of transistors QU , Q5 and Q6 connected 
 in the usual manner. The amplitude D/A converter is fed to the TTL-compatible 
 comparator where its output is compared against the output from the integra- 
 tor. Depending on whether the amplitude of the converter output is smaller 
 or greater than that of the integrator, the exponential current is diverted 
 to the integrator input either through diode Dl or through current mirror 
 Q1/Q2. Transistors Ql through Q5 belong to the CA3096 IC array. Upon reach- 
 ing an envelope breakpoint, the comparator begins to oscillate, causing the 
 2-bit counter to overflow and thereby generate a request pulse to the binary 
 sequence generator. When servicing the request, the sequence generator also 
 clears the counter, thereby providing proper initialization for the detection 
 of the next breakpoint. 
 
 With the circuit parameters shown, attack slew-rates cover a range from 
 1 V/ms to 1 V/s and decay slew-rates from 2 V/ms to 2 V/s. The one-octave 
 discrepancy between the two ranges is dictated by reasons of musical conven- 
 ience and is handled by offsetting the exponential range by a corresponding 
 amount in the course of decay ramps. The offsetting circuitry is centered 
 around resistor Rl and transistor switch Q.7. During an attack the comparator 
 output is at "0" and the offsetting circuitry is disabled by switch QT. Dur- 
 ing a decay the comparator opens switch QT, so that the current through re- 
 sistor Rl is directed to the base of exponential generator Qk , where it intro- 
 duces a voltage offset of about -18 mV. This causes the collector current 
 of QU to decrease by one octave, as desired. Trimmer R2 serves the purpose 
 of adjusting the base of the exponential range. 
 
65 
 
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 As will be seen in connection with musical parameter control, the appli- 
 cations of piece-wise linear function generators are not confined to sound 
 enveloping alone. When used in applications that do not require the one- 
 octave discrepancy between positive and negative slope ranges, the offset- 
 ting circuitry just described is simply omitted from the hardware realiza- 
 tion of the generator. 
 
67 
 
 V. PERCUSSION ENSEMBLES 
 
 Two of the four orchestras, in addition to the voice modules described 
 above, include also a l6-element percussion ensemble each. In principle, 
 percussion-like sounds can be synthesized by means of voice modules, as all 
 other sounds are. However, the fact that percussion sounds consist, to a 
 good approximation, of damped oscillations which are easy to produce elec- 
 tronically, makes it worthwhile to generate these sounds separately, by 
 means of dedicated hardware. This solution is intended to reduce the work- 
 load of the voice modules, so that they can be more efficiently allocated 
 to the synthesis of sounds of higher complexity. It also offers the benefit 
 of simpler control, as will appear below. 
 
 Figure 25 shows the basic percussion circuit configuration, which con- 
 sists of a trigger/envelope driver and a phase-shift oscillator. Normally 
 switch Ql is open and CI is charged at -15 V. The base network of Q2 is 
 such that, under these conditions, the circuit is unable to sustain oscil- 
 lation due to insufficient base bias to keep Q2 on. 
 
 The arrival of a trigger pulse causes Ql to close and discharge CI to 
 ground. The voltage step now appearing across R2 is large enough to bias Q2 
 in the active region, where the circuit is capable of sustained oscillations. 
 The frequency of oscillation is determined by the components included in the 
 feedback network of Q2. 
 
 After removal of the trigger pulse, Ql is opened and CI is allowed to 
 charge again to -15 V, at a rate determined by Rl , R2 and CI itself. This 
 voltage transition gradually turns Q2 off, via R2, and causes the amplitude 
 of oscillation to undergo a smooth decay to zero. The rate of decay is 
 determined by the rate at which CI charges to -15 V. Resistors R3 and Hk 
 
68 
 
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69 
 
 set the point at which the amplitude of oscillation reaches the zero value. 
 
 The trimmer shown is used for final tuning of the percussion pitch. 
 
 Thus, whenever the circuit is struck by a trigger pulse, it emits a 
 
 series of damped oscillations. By properly choosing the various component 
 
 values, the same basic configuration can be easily adapted to span an entire 
 
 family of percussion instruments like bass drums, tom-toms, bongos, claves , 
 
 wood-blocks, congas and castanets. As an example, the following are the 
 
 component values that have been used to synthesize a bongo: 
 
 Rl = 220 k CI = .05 yF 
 
 R2 = 1 M C2 = .0033 yF 
 
 R3 = 2.2 M C3 = .0033 yF 
 
 RU = 100 k Ck = .0033 yF 
 
 R5 = 150 k C5 = 750 pF 
 
 Figure 26 shows the simplified circuit schematic of an entire percussion 
 ensemble. The control portion involves two read/write memories addressed in 
 parallel by a variable -modulus counter. The information residing in the 
 lower memory specifies the pattern according to which the instruments are to 
 be struck in sequence. The information of the upper memory determines the 
 intensity with which each instrument is to be struck. 
 
 Circuit operation is rather straightforward. The arrival of a clock 
 pulse from the orchestra duration control steps the counter and causes two 
 new words of data to appear at the outputs of the respective memories. The 
 word from the upper memory, after D/A conversion, programs the attenuator for 
 the proper loudness. The word from the lower memory sets up the decoder so 
 that the 1 ms trigger pulse emitted by the one-shot is directed to the proper 
 instrument. Thus a sequence of clock pulses results in the generation of a 
 pattern of percussion sounds. Pattern length is determined by the modulus of 
 the counter, which can be programmed externally. 
 
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 Being provided with read/write capabilities, "both memories can be inde- 
 pendently updated to accommodate new patterns. This fact, together with the 
 programmability of the counter, makes the circuit suitable for generating a 
 wide variety of rhythms and patterns. Memory updating can be carried out 
 while the percussion ensemble is in action. To avoid synchronization prob- 
 lems, the leading edge of the duration clock is followed by the emission of 
 a sharp pulse by the 100 ns one-shot, which strobes the information waiting 
 at the memory data-inputs into the memory register being currently addressed. 
 
 The entire percussion ensemble is silenced by disabling the duration 
 clock. 
 
72 
 
 VI. SOUND DISTRIBUTION AND SWITCHING 
 
 6.1 Spatial Sound Distribution . 
 
 The four-channel output system discussed earlier is primarily intended 
 for the control of sound location in studio-type situations. For operation 
 in concert-hall environments, the output capabilities of the instrument are 
 expanded by the addition of 2k discrete channels, each complete with power 
 amplifying hardware to drive a corresponding loudspeaker. Spatial elements 
 are then controlled by routing sounds to the various loudspeakers which are 
 placed at different locations throughout the performance area. 
 
 This output configuration can handle spatial situations of much greater 
 complexity than the quadraphonic system. One example is offered by sound 
 traffic, an effect that can be readily produced by switching a sound from 
 one position to the next, along a chain of consecutive loudspeakers. By 
 permuting the order in which the loudspeakers are grouped into chains, one 
 can specify arbitrary paths of sound traffic. 
 
 Sound routing is handled by means of analog gates, or audio gates, as 
 they will more properly be termed henceforth. Being a two-state device, a 
 gate requires only one bit of control information, and this is a highly de- 
 sirable feature in that it maintains the overall information requirements 
 within manageable limits. Thus, a complete sound routing system necessi- 
 tates 2k audio gates, each gate being assigned to control the presence or 
 absence of sound on a corresponding channel. Accordingly, the specification 
 of an entire sound distribution pattern can be done with 2k bits of data. 
 
 To set a distribution pattern in motion, the associated information must 
 be properly processed, and a suitable device to perform this task is a cir- 
 cular shift-register. The register consists of 2k cells, each cell being 
 assigned to control the state of a corresponding audio gate. It is also 
 
73 
 
 provided with parallel load facilities for the direct entry of sound distri- 
 bution patterns. The order in which the gates are assigned to consecutive 
 register cells defines a corresponding chain of loudspeakers extending 
 throughout the performance space. Thus, shifting a bit pattern around the 
 register causes an identical sound pattern to move around the loudspeaker 
 chain. The direction of motion is determined by the direction of shift. 
 
 To offer a wider choice of sound traffic paths, the shift-register is 
 broken down into six shorter registers, as shown in Figure 27. More complex 
 sound paths are then specified by permuting the order in which the shorter 
 registers are joined together. Permutation patterns are programmed by means 
 of a 6 x 6 interconnection matrix. When the (i,j)-th bit of the matrix is 
 at logic one, the serial output from the i-th register is diverted to the 
 serial input of the j-th register. Thus each register can be connected to 
 any other register, including itself. 
 
 Although this scheme does not exhaust all possible traffic paths that 
 can be specified with a system of 2H loudspeakers, it offers enough variety 
 to satisfy a wide range of musical needs without rendering control unclear 
 or too complex. 
 
 To preserve its spatial individuality, each of the four orchestras is 
 provided with an independent sound distribution system, so that it can con- 
 trol the location and movement of its sounds according to its own patterns 
 and rates.* A block diagram of the entire sound distribution system is 
 shown in Figure 28. The system has four input busses, each of which carries 
 the sounds pertaining to one of the four orchestras. These busses are driven 
 by the quadraphonic output system discussed earlier. The assignment of each 
 orchestra to one of the quadraphonic outputs is accomplished by proper 
 
 * The sound distribution control system was designed by J. Divilbiss 
 and R. Borovec. 
 

 
 
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76 
 
 initialization of the directional control inputs to the various attenuator/ 
 locator devices. 
 
 6.2 Audio Switching . 
 
 As shown in the block diagram of Figure 28, the realization of the sound 
 distribution system requires a total of 96 audio gates. With a number of this 
 magnitude, the cost per gate plays an important role in the choice of the cir- 
 cuit realization to be adopted. 
 
 An audio gate, besides satisfying such obvious requirements as low dis- 
 tortion, high on-to-off transfer ratio, and low control signal feedthrough, 
 must also exhibit specific transient characteristics in order to ensure pro- 
 per audio switching. The last requirement is motivated by the fact that 
 when a sound is switched on or off, spurious partials are generated which 
 may alter the tonal character of the sound considerably, as has been discussed 
 in connection with sound enveloping. 
 
 As Fourier analysis reveals, the amount of spurious partials accompany- 
 ing sound switching usually increases with the rate at which sound builds up 
 or decays. Thus, unwanted switching effects can be easily reduced by employ- 
 ing gates with low switching speeds. From the viewpoint of sound movement 
 control, however, it is desirable to have fast gates so that sounds can be 
 switched around the performance space at arbitrary rates. A compromise be- 
 tween the two conflicting requirements can be determined experimentally. 
 
 Analog gates based on solid-state, electro-optical devices like Ray- 
 si stors or Vactrols usually exhibit predetermined and highly asymmetrical 
 switching characteristics which render these components unsuited to the present 
 application. The cost of these devices at the time of design was also a key 
 factor that contributed to their rejection. Analog gates of the FET or MOS 
 
77 
 
 type must be ruled out "because of their high switching speeds and the lack of 
 a simple and inexpensive circuit configuration to slow them down. 
 
 The specifications set forward above have been met with variable-trans- 
 conductance transistor pairs. The switching characteristics of these de- 
 vices are set to their prescribed values by means of simple RC networks, as 
 shown in Figure 29« To illustrate circuit operation, consider the first 
 gate, which consists of transistor pair Q1/Q2 and biasing transistor QU. 
 The gain or transmission of the gate is determined by the collector current 
 of QU. Since Qh is connected as an emitter follower, its collector current 
 and, hence, the transmission of the gate, is modulated by the voltage applied 
 to the base of QU . This voltage is in turn controlled by current switch Q3. 
 When the control input is high, Q3 is on and its collector current develops 
 a voltage drop across D2 and R2 in series, which biases Q^ on also. When the 
 control input is low, transistor Q3 is off, and this causes QU to be off, due 
 to the absence of collector current from Q3 to keep the base network of Q4 
 biased. 
 
 Although Q3 is switched from one state to the other at logic speed, the 
 voltage transitions at the base of QU are slowed down by damping capacitor C. 
 Thus, gate transmission builds up and decays according to the transient char- 
 acteristics of an RC element. The switching time-constant equals (R2 + r_) • 
 C ~ 12 ms , where r is the dynamic resistance of diode Dl. This time-constant 
 value represents a satisfactory compromise between the need for fast sound 
 movement and that for spurious partials reduction. 
 
 Diode Dl compensates for the presence of the base-to-emitter voltage 
 drop of Qk. Hence it prevents the collector current of Qk from falling too 
 rapidly to zero when the voltage at the base of QU decays below the value of 
 
78 
 
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 cfl 
 
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79 
 
 a junction voltage drop. Trimmer Rl is used to adjust for zero dc offset at 
 the gate output. 
 
 The outputs from the four gates are mixed together by tying correspond- 
 ing collectors of the transistor pairs in parallel. The circuit realization 
 of a quadruple audio gate requires two CA305^ transistor arrays, as shown. 
 
 The balanced outputs from the gates are buffered to the subsequent stage 
 by common-base transistors Q5 and Q6. The purpose of these transistors is to 
 keep the collectors of the gates near zero potential and, therefore, minimize 
 the power dissipated by the trans conductance pairs. This precaution is taken 
 to reduce temperature variations of the dc output offset which constitutes 
 the primary cause of control signal feedthrough. Indeed, when a gate is 
 switched on or off, its dc offset is switched on or off as well, resulting 
 in a form of coupling of the control input to the output. Control feed- 
 through is particularly noticeable when the distribution controller is pro- 
 grammed for fast sound movement and loudness control is, at the same time, 
 programmed for a rest. This problem does not arise in connection with the 
 programmable attenuator/locator because, whenever the attenuator is programmed 
 for a rest, the transconductance pairs of the locator are automatically off. 
 With the precaution illustrated above, control signal feedthrough is main- 
 tained more than 60 db below full signal level. 
 
 The task of converting the balanced outputs from the buffers into a 
 single-ended signal suitable to drive an 8fi-loudspeaker is accomplished by 
 the circuit of Figure 30. The circuit consists of the usual differential 
 amplifier and a power stage to boost the amplifier drive capability. The 
 trimmer shown is used to adjust the quiescent current of the power stage 
 for the prevention of thermal runaway. Power stage and loudspeaker are dc 
 
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81 
 
 coupled to prevent the use of a "bulky decoupling capacitor. Two lA-fuses 
 are used to protect the loudspeaker against amplifier malfunction. 
 
 The circuits of Figures 29 and 30 are easily accommodated on a single 
 printed-circuit board of standard size (h 1/2" x 6"). 
 
82 
 
 VII. CONTROL SYSTEM 
 
 7.1 Musical Parameter Control . 
 
 The material presented so far has dealt primarily with circuits and 
 techniques for the synthesis and processing of sounds. To make musical para- 
 meter control possible, all modules encountered respond to some form of 
 digital or analog control which allows their electronic characteristics to be 
 programmed externally. We shall now examine in greater detail how these cir- 
 cuit parameters are actually programmed in order to produce music. 
 
 As stated at the beginning of this work, all control functions are speci- 
 fied in purely digital form. Several of the sound modules encountered, namely, 
 equitempered pitch generators, duration controllers, envelope generators and 
 digital waveshape generators, are already equipped with digitally programmable 
 inputs. All that needs to be done, then, is to feed these control inputs with 
 a stream of binary information of the proper word length. Most of the para- 
 meters of musical interest, however, are of a continuous, rather than a dis- 
 crete nature, as reflected by the fact that the corresponding circuit char- 
 acteristics are controlled by analog voltages instead of digital signals. To 
 make these parameters accessible to digital control, analog sound modules 
 must be preceded by a proper digital-to-analog interface. This task is 
 accomplished by means of piece-wise linear function generators of the same 
 type as those used in the generation of envelopes. Although these devices 
 were discussed in connection with loudness control because that is probably 
 their most illustrative application, piece-wise function generators are of 
 far more general applicability, in that they can be used to control the time 
 evolution of any analog parameter. 
 
 This generalized notion of envelope for the uniform control of all con- 
 tinuous quantities results in a high degree of control as well as hardware 
 
83 
 
 standardization. The use of this type of interface to control analog 
 modules also results in a significant data-rate reduction, since the para- 
 meters to be controlled are the envelope breakpoints. Indeed, had the inter- 
 face consisted of straight D/A converters, the quantities to be specified 
 would have been individual envelope samples, a task that involves a much 
 higher data-rate than the mere specification of breakpoints. 
 
 In the light of the above discussion, all sound modules may now be 
 treated as programmable devices that respond to purely digital control. 
 Regardless of their nature, significance and word-length, the dynamic con- 
 trol of musical parameters is accomplished in a uniform manner by means of 
 binary sequences. As implied by its name, a control sequence consists of a 
 series of binary words which are successively spaced in time at determinate, 
 but not necessarily uniform, time intervals. This control approach is in- 
 tended to be of enough generality as to leave the programming of sound 
 modules open to a broad choice of digital systems and techniques for its 
 implementation . 
 
 Binary sequences are generated by programmable word generators and by 
 the human operator. Depending on the structural level at which they affect 
 music, control sequences are classified as micro structural or macrostructural. 
 
 Micro structural sequences are those effecting individual parameter 
 changes and, hence, are the sequences that control the characteristics of 
 sound modules directly, as shown in Figure 31. Since perceptual parameters 
 change at rates in the range of 1 to 100 Hz, these sequences are too fast for 
 human control in real-time. Hence microstructural sequences are generated by 
 word generators that are built into the system. 
 
 Macrostructural sequences are used to manipulate microstructural se- 
 quences for the creation of more complex musical structures. To make this 
 
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 manipulation possible, all micro structural sequence generators are them- 
 selves externally programmable. Since they affect music at a higher level 
 of structural complexity, macro structural sequences are characterized by 
 lower timing rates, which may typically be of the order of .01 to 1 Hz. 
 These are rates at which a human operator can comfortably perform control 
 actions in real-time. Hence, the generation of macrostructural sequences for 
 the control of microstructural sequence generators is made accessible to the 
 operator. 
 
 If so desired, the operator can delegate the generation of macrostruc- 
 tural sequences to the system itself. To handle this task, the latter is 
 also provided with macrostructural sequence generators, in addition to the 
 microstructural ones already discussed. The information associated with 
 these macrostructural sequences is called automatic, as opposed to that 
 generated by the operator which is called manual. The specification of 
 whether macrostructural information is to be generated manually or automat- 
 ically is expressed by the operator himself in terms of one-bit sequences to 
 control the information steering device, which consists of data selectors. 
 To allow higher levels of control complexity, macrostructural sequence gen- 
 erators are also programmable. However, their programming is done by the 
 operator exclusively. 
 
 7.2 Programmable Binary Sequence Generation . 
 
 A device that has proved to be particularly attractive for the genera- 
 tion of binary control sequences, both in terms of simplicity and cost, is 
 the feedback shift -register . As shown in Figure 32 for the case of U-bit 
 word sequences, this device consists of a shift-register and a combinational 
 circuit which computes the next value of the serial input as a function of 
 
86 
 
 PARALLEL OUTPUTS 
 
 
 
 
 r 
 
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 A 
 
 
 
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 A 
 
 
 FEEDBACK 
 FUNCTION 
 
 I 
 
 
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 S 
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 Figure 32. Feedback Shift-register. 
 
 
 Pi 
 
 VRALLEL OUTPU' 
 
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 SEQUENCE 
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87 
 
 the present state of the register. At every shift-pulse the register under- 
 goes a transition from one state to another, so that a series of shifts car- 
 ries the register through an entire sequence of states. Thus, by using a 
 shift-register with parallel output facilities, the sequence of words 
 appearing at the output lines can be used to control musical parameters 
 directly. 
 
 Shift-register sequences have been widely investigated in the litera- 
 ture and will not be discussed in detail here. It may suffice to point 
 out that, in general, with a given feedback function, a register can gen- 
 erate a whole family of different sequences, and that different feedback 
 functions yield different families of sequences. Thus, the versatility of 
 a feedback shift-register can be considerably expanded if a programmable 
 rather than a fixed feedback function is used. 
 
 Once a particular feedback function has been programmed, the register 
 can be initialized to any one of the sequences of the corresponding family 
 by loading it with a word of that sequence. To make this possible, the 
 register is provided with parallel-load facilities. The use of a bidirec- 
 tional shift-register also increases the versatility of this device, since 
 the sequences resulting from left shifts are generally different from those 
 obtained with right shifts. 
 
 Feedback shift-register sequences enjoy the musically attractive fea- 
 ture of being periodic. This means that after a sufficient number of shifts, 
 a sequence repeats itself over and over again, until the system is programmed 
 for a different sequence. 
 
 A convenient way of implementing a programmable feedback function is by 
 
 means of a modulo-2 adder with programmable patterns of active inputs. It 
 
 27 
 can be easily shown that the periodic lengths of the sequences obtained 
 
88 
 
 with this class of feedback functions are bounded by 2 -1, where n is the 
 shift-register length. Figure 33 shows the realization of a U-bit word se- 
 quence generator based on the above ideas. 
 
 By exploiting the options made available by the various control inputs, 
 the same device can be programmed to generate a broad class of control se- 
 quences. In general, each sequence posesses a musical significance of its 
 own. Hence, the generator offers a set of basic compositional elements which 
 can be easily manipulated by the improviser to create more complex musical 
 structures. 
 
 In order to satisfy a class of musical needs as broad as possible, it 
 is necessary that musical parameters be programmable over wide dynamic ranges. 
 However, in defining a particular musical context, the composer is likely to 
 use only limited portions of the dynamic ranges available. Thus, control se- 
 quences can be musically useful only if the associated stream of information 
 can be constrained within appropriate limits or windows. 
 
 Since feedback shift-registers do not offer facilities for information 
 biasing as such, this task must be handled independently. A simple way to 
 achieve this goal is to partition the bits of a control word into a most- 
 significant and a least-significant group, and to handle the two groups 
 separately, the first with a macrostructural sequence and the second with a 
 microstructural sequence. This causes the resulting information to be biased 
 within windows whose width is determined by the length of the least-signifi- 
 cant portion of each word, and whose position within the total range is con- 
 trolled by the most-significant data. 
 
 Although adequate for the control of certain parameters, this scheme is 
 not sophisticated enough for others, due to the fact that windows are con- 
 strained to be of fixed widths. 
 
89 
 
 A more flexible scheme of information biasing relies on programmable 
 window detectors, as shown in Figure 3^ for the case of 4-bit words. The 
 detector output results in an assertion only if the input information falls 
 within the limits prescribed at the control inputs. Hence, this device can 
 readily be incorporated into an information-filtering scheme for the biasing 
 of entire control sequences within arbitrarily programmable windows. 
 
 Since sequence generators that are equipped with all the programming 
 facilities illustrated above possess a substantial number of control inputs, 
 only a centralized set of devices of this type is used. The related se- 
 quences are then transferred to local memories, where they are recirculated 
 for dynamic parameter control. These peripheral memories consist of read/ 
 write IC memories addressed by programmable-modulus counters, in the manner 
 illustrated in connection with the control of percussion sounds. 
 
90 
 
 Xl> 
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 Ul U2 U3 U4 
 V V V V 
 
 Bl B2 B3 B4 
 
 7485 A>B 
 
 Al A2 A3 A4 
 
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 7485 A<B 
 
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 t> 
 
 w 
 
 w = 1 
 
 L < X < U 
 
 Figure 3^. Digital Window Detector. 
 
91 
 
 VIII. CONCLUSION 
 
 A hardware system has "been presented which allows for the composition 
 of electronic music in real-time. Real-time operation is achieved by a 
 proper functional partition which assigns analog hardware to the synthesis 
 and processing of sounds and digital hardware to the control of musical 
 parameters. Throughout the design of the system particular emphasis has 
 been placed on assuring wide ranges of sound quality as well as flexibility 
 of parameter control. 
 
 The use of sophisticated hardware made available by recent technology 
 has led to the development of sound modules that afford a much richer range 
 of tonal and instrumental possibilities than obtainable from present-day 
 analog synthesizers. Furthermore, the output configuration based on a system 
 of loudspeakers distributed throughout the performance space offers a rela- 
 tively new approach to the spatial characterization of sounds and ensures a 
 total immersion and involvement of the listener with the sound/space environ- 
 ment. 
 
 The adoption of generalized envelope functions for the dynamic control 
 of continuous musical quantities introduces a high degree of control uni- 
 formity, in that all parameters, be they discrete or continuous, can be 
 handled in the same manner. Parameter control is based on the notion of dis- 
 crete control sequences, which are characterized by clocking rates of about 
 100 Hz or less. While involving a relatively low data-rate, this approach 
 is also of enough generality as not to put definite constraints on the type 
 of digital systems to be used for the control of sound modules. 
 
 The structural nature of music allows a correspondingly structured orga- 
 nization of control which is particularly significant in the real-time 
 
92 
 
 interaction between composer/performer and instrument. In this respect, a 
 general distinction between micro structural and macrostructural control se- 
 quences has been emphasized so that a proper balance between operator re- 
 sponsibilities and machine responsibilities can be achieved. 
 
 The fact that technology continues to market analog and digital systems 
 of increasing complexity and competitive cost is bound to have a significant 
 effect on electronic music. For instance, with the recent appearance of 
 microcomputers, it is now becoming economically feasible to use one or more 
 
 of these devices to program sound modules, so that the sophistication of the 
 
 28 
 control system outlined in this work can be expanded further. Technology 
 
 advances are bound to affect also sound processing hardware and techniques. 
 In this respect, it is highly desirable that sound modules become more digi- 
 tally oriented, since digital systems enjoy the intrinsic property of being 
 far more accurate and stable than analog circuits. In the light of these 
 remarks , it seems appropriate to speculate that the electronic music systems 
 of the next generation will be based on a central, general -purpose computer 
 controlling a collection of peripheral, special-purpose computers whose task 
 is the synthesis and processing of sounds. 
 
93 
 
 LIST OF REFERENCES 
 
 1. Moore, F. R. , "Computer Controlled Analog Synthesizers," Bell Telephone 
 
 Laboratories, Inc. (internal Memo.). 
 
 2. Moog, R. A., "Voltage Controlled Electronic Music Modules," J. Audio 
 
 Engineering Society, Vol. 13, p. 200, 1965). 
 
 3. Moog, R. A., "Electronic Music: Its Composition and Performance," 
 
 Electronics World, February 1967 • 
 
 h. Friend, D. , "The ARP Synthesizer: A New Instrument for Musical Com- 
 position and Performance," AES Preprint #7^2 (B-3), Audio Engineering 
 Society 39th Convention, October 12-15, 1970. 
 
 5. Mayer, A., "The PUTNEY: A New Generation of Synthesizers," AES Preprint 
 
 #750 (B-l), Audio Engineering Society 39th Convention, October 12- 
 15, 1970. 
 
 6. Mathews, M. V., The Technology of Computer Music (MIT Press, Cambridge, 
 
 Mass. 1969). 
 
 7. Mathews, M. V., F. R. Moore and J. C. Risset, "Technology and Future 
 
 Music," Bell Telephone Laboratories, Inc. (internal Memo.) 
 
 8. Beauchamp, J. W. , "Electronic Music Equipment: 1950-1970," Experimental 
 
 Music Studio Technical Report No. 21, University of Illinois, 
 February 1970. 
 
 9. Risset, J. C. and M. V. Mathews, "Analysis of Musical Instrument Tones," 
 
 Physics Today, Vol. 22, No. 2, pp. 23-30, 1969. 
 
 10. Chowning, J. M. , "The Synthesis of Complex Audio Spectra by Means of 
 
 Frequency Modulation," J. Audio Engineering Society, Vol. 21, 
 pp. 526-53^, September 1973. 
 
 11. Gibbons, J. F. and H. S. Horn, "A Circuit with Logarithmic Transfer 
 
 Response over 9 Decades," IEEE Trans. Circuit Theory, Vol. CT-11, 
 pp. 378-381+, September 196*+. 
 
 12. Borlase , W. and E. David, "Log Circuits," Analog Devices, Inc., Applica- 
 
 tion Note, March 1969- 
 
 13. Gilbert, B. , "A Precise Four Quadrant Multiplier with Subnanosecond Re- 
 
 sponse," IEEE J. Solid-State Circuits, Vol. SC-3, pp. 365-373, 
 December 1968. 
 
 lU. Gilbert, B. , "Comment on 'Single-Ended-Input Single-Ended-Output Four- 
 Quadrant Analog Multiplier'," IEEE J. Solid-State Circuits, Vol. SC-7, 
 p. U3^, October 1972. 
 
 15. Mitra, S. K. (ed.), "Active Inductorless Filter," IEEE Press, Inc., 
 New York, 1971. 
 
9h 
 
 16. Capella, D. , "FETs as Voltage-Controlled Resistors," Siliconix, Inc., 
 Application Note, February 1973. 
 
 IT- Bruton, L. T. , "Electronically Tunable Analog Active Filters," IEEE 
 .Trans. Circuit Theory, Vol. CT-19, pp. 299-301, May 1972. 
 
 18. Sparkes, R. G. and A. S. Sedra, "Programmable Active Filters," IEEE 
 
 J. Solid-State Circuits, Vol. SC-8, pp. 93-95, February 1973. 
 
 19. Kaehler , J. A., "Periodic-Switched Filter Networks - A Means of Ampli- 
 
 fying and Varying Transfer Functions," IEEE J. Solid-State Circuits, 
 Vol. SC-U, pp. 225-230, August 1969. 
 
 20. Bruton, L. T. and R. T. Pederson, "Electronically Tunable Time-Multi- 
 
 plexed Active Filters," IEEE J. Solid-State Circuits, Vol. SC-7, 
 pp. 266-267, June 1972. 
 
 21. Kerwin, W. J., L. P. Huelsman, and R. ¥. Newcomb , "State-Variable Syn- 
 
 thesis for Insensitive Integrated Circuit Transfer Functions," 
 IEEE J. Solid-State Circuits, Vol. SC-2, pp. 87-92, September 1967. 
 
 22. Thomas, L. C. , "The Biquad: Part I - Some Practical Design Considera- 
 
 tions," IEEE Trans. Circuit Theory, Vol. CT-18, pp. 350-1*57, 
 May 1971. 
 
 23. Geffe, P. R. , "Toward High Stability in Active Filters," IEEE Spectrum, 
 
 Vol. 7, pp. 63-66, May 1970. 
 
 2k. Oneil, B. , "A Precision Four Quadrant Multiplier," Intersil, Inc., 
 Application Bulletin A011, June 1972. 
 
 25. Wilson, G. R. , "A Monolithic Junction FET-NPN Operational Amplifier," 
 
 1968 Internat'l. Solid-State Circuits Conf . , Digest of Tech. Papers, 
 pp. 20-21. 
 
 26. Wittlinger, H. A. , "Applications of the CA3080 High-Performance Opera- 
 
 tional Trans conductance Amplifiers," RCA's Linear Integrated Circuits 
 Application Note ICAN-6668, March 1972. 
 
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 1967. 
 
 28. Franco, S. , "The Oberlin Hybrid Project," Oberlin Conservatory of Music 
 
 Internal Memo, June 197*+. 
 
95 
 
 VITA 
 
 Sergio Franco was born in Capriva del Friuli , a rural village of north- 
 eastern Italy. After graduation in Physics from the University of Rome, he 
 taught high school Mathematics and Physics in Rome. In 1967 he was awarded 
 a FuTbright Fellowship to continue his studies in the USA. He first joined 
 the Department of Physics of Clark University, Worcester, Massachusetts, as 
 a teaching assistant, where he earned a M.A. degree in Physics in the area 
 of the Mossbauer effect. In 1968 and 1969 he was associated with the Illi- 
 ac III Project at the Department of Computer Science of the University of 
 Illinois at Urbana, where he designed the analog circuitry of the Scan/Dis- 
 play System. After attending graduate courses in Computer Science at the 
 University of Toronto during the years 1970 and 1971? he returned to the 
 Urbana campus, where he became associated with the Coordinated Science 
 Laboratory and the School of Music. Since then, his main interest has been 
 in the area of electronic music. During 1971 and 1972 he worked with the 
 composer, Salvatore Martirano, in the development of the musical system de- 
 scribed in this thesis. In 1973 he joined the Oberlin Conservatory of 
 Music where he teaches courses in analog and digital design of Electronic 
 Music Systems, and is currently developing the Oberlin Hybrid Project, a 
 computer controlled system for the composition of music in real-time. 
 Mr. Franco has published a number of articles in Physics and in Electronics. 
 
BIBLIOGRAPHIC DATA 
 SHEET 
 
 1. Report No. 
 
 UIUCDCS-R-7U-677 
 
 3. Recipient's Accession No. 
 
 4. Title .ind Subtitle 
 
 HARDWARE DESIGN OF A REAL-TIME MUSICAL SYSTEM 
 
 5. Report Date 
 
 December 197^ 
 
 7. \uth<>r(si 
 
 Sergio Franco 
 
 8. Performing Organization Kepi 
 
 No - UIUCDCS-R-7 i +-677 
 
 •r forming Organization Name and Address 
 
 Department of Computer Science 
 
 University of Illinois at Urbana-Champaign 
 
 Urbana, Illinois 6l801 
 
 10. Project/Task/Work Unit No. 
 
 11. Contract /Grant No. 
 
 I 
 
 rsoring Organization Name and Address 
 
 implement ary Notes 
 
 13. Type of Report & Period 
 Covered 
 
 technical 
 
 14. 
 
 lift 
 
 This thesis describes the hardware design of a hybrid system for the composition 
 j and performance of electronic music in real-time. While analog circuitry is primarily 
 i employed in the generation and processing of sounds, digital circuitry is devoted to 
 i the exercise of control, under the immediate supervision of the composer/performer, 
 j Such a digital vs. analog partition, together with a proper choice of the man vs. 
 j machine interface, is intended to satisfy the much emphasized musical need for the 
 j immediate, real-time interaction between the composer/performer and his instrument. 
 
 17. Key Words and Document Analysis. 17a. Descriptors 
 
 Electronic Music 
 
 Real-Time Music Synthesis 
 
 Musical Waveshape Systhesis/Control 
 
 [17b Hv!it the rs 'Opcn-hnded Terms 
 
 I 
 
 17c I OSATI Field/Group 
 
 18 W ail ability Statement 
 
 unlimited 
 
 19. Security Class (This 
 Report) 
 
 UNCLASSIFIED 
 
 20. Security Class (This 
 Page 
 
 UNCLASSIFIED 
 
 21. No. of Pag< 
 
 100 
 
 22. Price 
 
 $8.00 ei 
 
 3 for $22.00 
 
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