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 UILU-ENG 77 1701 
 
 TRANSMISSION OF ANALOG SIGNALS USING BURST TECHNIQUES 
 
 by 
 MARTIN WOLFF 
 
 January 1977 
 
 DEPARTMENT OF COMPUTER SCIENCE 
 
 UNIVERSITY OF ILLINOIS AT URBANA-CHAMPAIGN • URBANA, ILLINOIS 
 
 University ot Illinois 
 t Urbana-Cham urn 
 
UIUCDCS-R-77 -838 
 
 TRANSMISSION OF ANALOG SIGNALS 
 USING BURST TECHNIQUES 
 
 BY 
 MARTIN WOLFF 
 
 January 1977 
 
 Department of Computer Science 
 University of Illinois 
 Urbana, Illinois 61801 
 
 This work was supported in part by Contract No. N00014-75-C-0982 and 
 the Department of Computer Science at the University of Illinois, Urbana- 
 Champaign, and was submitted in partial fulfillment of the requirements 
 for the degree of Master of Science in Electrical Engineering, at the 
 University of Illinois. 
 
Ill 
 
 ACKNOWLEDGMENT 
 
 The author wishes to thank his thesis advisor Professor W. J. 
 Poppelbaum for the opportunity to be a member of the Information Engineer- 
 ing Laboratory and for his support and guidance throughout the term of 
 the project. He also wishes to thank all the members of the Information 
 Engineering Laboratory for their continual friendship and assistance and 
 their willingness to allow him to appropriate the 350 MHz oscilloscope. 
 Thanks are due to Mrs. Kathy Gee for typing the thesis, Mr. Stan Zundo 
 for doing the drawings, and Mr. Frank Serio and his staff for their help 
 in putting the project together. Finally he would like to offer the 
 classical thanks to his wife Lois for her support and encouragement. 
 
IV 
 
 TABLE OF CONTENTS 
 
 Page 
 
 1. INTRODUCTION 1 
 
 2. BURST ENCODING 2 
 
 2.0 Burst Basics 2 
 
 2.1 Encoding Using Burst Techniques 3 
 
 3. BURSTCOMM I 12 
 
 3.0 Burstcomm I Description 12 
 
 3.1 Burstcomm I Results 16 
 
 4. BURSTCOMM II 18 
 
 4.0 Burstcomm II Description 18 
 
 4.1 Audio and Sensor Circuits 20 
 
 4.2 Video Circuits 23 
 
 4.3 Optical Link Circuits 24 
 
 4.4 Burstcomm II Results 26 
 
 5. BIBURST 28 
 
 6. CONCLUSIONS 30 
 
 APPENDIX 
 
 CIRCUIT AND TIMING DIAGRAMS 31 
 
 LIST OF REFERENCES 42 
 
1. INTRODUCTION 
 
 The fundamental idea of burst processing is to combine the 
 advantages of stochastic processing (utter simplicity) with the advantages 
 of weighted binary (higher precision with reasonable bandwidth) to obtain 
 a system lying in the middle ground. The result is a system which allows 
 continous processing of non-synchronized signals yielding low precision 
 immediate results and higher precision final results through averaging. 
 
 The simplicity of circuitry and minimization of synchronization 
 requirements makes burst processing potentially useful in the area of 
 communications. This paper then proposes to investigate various ways in 
 which burst techniques can be used in communications and to give the 
 design considerations and results of three demonstration systems. 
 
2. BURST ENCODING 
 2.0 Burst Basics 
 
 A basic building block in burst processing is the Block Sum Registe 
 (see Figure 1). The voltage sum represents an average of the contents of 
 the shift register. The BSR therefore can act as an integrator on the 
 input bit stream. If each group on N bits represents a complete sample 
 of an analog signal, the BSR does indeed function as a perfect integrator. 
 Standard compacted bursts have this property and can be decoded with 
 a simple BSR. 
 
 Modifying the BSR allows various other encoding schemes to be 
 used. With bidirectional BSRs, a form of delta modulation can be used. 
 Companding becomes easily realizable in this case with weighted non-equal 
 current sources. 
 
 CLOCK 
 
 CLEAR 
 
 INPUT- 
 
 N BIT SHIFT REGISTER 
 
 N EQUAL CURRENT SOURCES 
 
 i 
 
 ■►OUTPUT 
 
 
 VOLTAGE SUM 
 
 Figure 1. BJock Sum Register 
 
2. 1 Encoding Using Burst Techniques 
 
 The burst encoder in the original proposal is the ramp encoder 
 (Figure 2a) which consists of a BSR connected as a stairstep generator and 
 a comparator. The comparator compares the stairstep to an analog input 
 generating compacted bursts, i.e. bursts where all ones or all zeros are 
 contiguous within a sample. The basic ramp encoder has the disadvantage 
 of requiring a relatively high sample bit rate. Because the stairstep 
 samples the input at a time dependent on its voltage level, the input 
 
 must change less than one level of quantization during one stairstep 
 
 3 
 to prevent additional noise from being introduced. Specifically, 
 
 a lVpp sine wave of frequency f with maximum slope around t = must 
 
 vary less than one level of quantization 1/N (for N bit bursts and a 1 
 
 volt stairstep). Therefore for bit rate f (t = + l/2f ), 
 
 N27rf -N2uf 
 1/N > .5 (sin-p-^- si n 2f c ) (D 
 
 s s 
 
 or where F is the normalized sampling frequency f s /f~ 
 
 F s > Ntt/ARCSIN(1/N) (2) 
 
 For large values of N x F $ equation 2 becomes 
 
 F $ > N 2 tt (3) 
 
 Note that all transmissions discussed in this thesis (with 
 the exception of 2e - BIBURST) are non return-to-zero. This implementa- 
 tion is different from transmissions in previous papers on Burst Process- 
 ing which are all return-to-zero. Given external clocking, the non return- 
 to-zero system would be used in real systems. 
 
 With the addition of sample/hold circuitry (Figure 2b) the bit 
 rate can be reduced to the level of the Nyquist sampling criterion, i.e. 
 samples at twice the signal frequency, 
 
 F s > 2 N (4) 
 
CLOCK CLEAR 
 
 INPUT © 
 
 COMPACTED 
 BURSTS 
 
 Figure 2a, 
 
 Ramp Encoder 
 
 CLOCK 
 
 CLEAR 
 
 INPUT © 
 
 COMPACTED 
 BURSTS 
 
 Figure 2b. Improved Ramp Encoder 
 
 CLOCK 
 
 INPUT © 
 
 UNCOMPACTED 
 BURSTS 
 
 Figure 2c Delta Block Encoder 
 
CLOCK 
 
 
 SL - '0 
 
 MODE «.« ', 
 SR - 1 
 
 i 
 
 
 1 
 
 1 
 
 s 
 
 
 ■»'— 
 
 BIDIRECTIONAL 
 
 BSR 
 
 
 
 
 
 [>- 
 
 
 
 
 UNCOMPACTED 
 
 INPU1 
 
 r n 
 
 X 
 
 BURSTS 
 
 
 
 
 Figure 2d. Improved Delta Block Encoder 
 
 SL 
 
 CLOCK 
 
 i 
 
 SR 
 
 BIDIRECTIONAL 
 
 BSR 
 
 8 = 7 STEP 
 
 INPUT© ii 
 
 •o' 
 
 8 
 
 -1 + 
 
 8 
 
 + i - 
 
 :=Q 
 
 3 LEVEL 
 OUTPUT 
 
 H - SR 
 
 GND- HOLD 
 
 L - SL 
 
 Figure 2e. Biburst Encoder 
 
A different encoder using the same basic components as the 
 ramp encoder is the delta block encoder (Figure 2c). The BSR tracks 
 the input signal by clocking in ones or zeros depending on whether it is 
 below or above the input. Again, as in the basic ramp encoder, the burst 
 bit rate must be sufficiently high so that the input changes less than 
 one level of quantization per sample. Since a new sample is effectively 
 taken with each bit, it would seem that 
 
 F s > tt/ARCSIN(1/N) (5) 
 
 or for large N x F 
 
 F s > ttN (6) 
 
 is sufficient. Unfortunately, the BSR only changes its value if the 
 input bit is the opposite of the final bit. If the bit falling off is 
 the same as the bit coming in the BSR does not change. Since the input 
 signifies a desired change, this error must be eliminated. The tracking of 
 the input must be made independent of the history of the signal. 
 
 Figure 2d using bidirectional shift registers is one solution. 
 An increase of the BSR is guaranteed regardless of the state of the BSR 
 (assuming of course the BSR is not already full of ones or zeros). Equa- 
 tion 5 is now valid for a single frequency input. 
 
 For a bandlimited input, obtaining a slope overload relation 
 becomes more difficult. Extensive work has been done analyzing noise in 
 conventional delta-modulation systems which the system in Figure 2d 
 closely resembles. The major difference is that using a BSR as an 
 integrator gives a maximum and a minimum output (and hence input if the 
 BSR is to track the whole input) DC voltage. The theoretical analyses 
 of conventional delta-mod have assumed no absolute maximum DC input 
 voltage, instead they have dealt with an input of a certain rms power. 
 
A straightforward method of determining a slope overload criterion 
 for the BSR system will be given after a short review of some useful 
 results in the literature. 
 
 Noise in delta-mod systems can be grouped into two categories; 
 slope overload noise or noise caused by failure to track the input in the 
 regions of maximum slope, and granular or quantization noise inherent 
 in quantizing an analog signal. Since the input is not excursion voltage 
 
 limited, any input slope is possible for a bandlimited signal and slope 
 
 5 
 overload results must be expressed in terms of probabilities. O'Neal 
 
 and Protonotarious have obtained results for slope overload noise with 
 
 good agreement with experimental results. Using curves they have calculated, 
 
 one can determine an optimal step size given a sampling frequency. Below 
 
 this step size slope overload noise predominates and above the step size 
 
 granular noise increases. For a RC limited gaussian input and F = 64 
 
 for example, the optimum step size for a IV input is approximately .047. 
 
 rr 
 
 This corresponds to a 21 bit BSR in a BSR system even though the calcula- 
 tions are not exactly correct for the DC maximum limited BSR system. 
 To totally eliminate slope overload noise, a frequently used 
 
 criterion is to set the maximum slew rate of the encoder to four times 
 
 7 8 
 the rms slope of the input. ' For gaussian signals, the probability 
 
 that slope overload error can occur under these conditions is less than 
 
 -5 9 
 4 x 10 . Expressing this relation for a IV system with step size 
 
 rr 
 
 1/N and F s = f s /f c where f c is the cutoff of the gaussian input, 
 
 F s > 4tt N//6 (7) 
 
 5 7 
 
 Granular noise has been examined rigorously * but a 
 
 crude method exists which gives quite accurate results. The noise 
 spectrum of granular noise is assumed to be white over a bandwidth equal 
 to f s . Given that noise error can vary from -1/N to +1/N at any time 
 
with equal probability, total average noise power is equal to 
 
 N , 1/N x 2 dx - 1/(3N 2 ). (8) 
 
 2 M/n 
 
 The non-filterable portion of this (frequencies within the signal band- 
 width) is clearly the total power x f c /f s - Therefore, for the non- 
 filterable noise power N , 
 
 N g = 1/(N 2 F S 2 3) (9) 
 
 Since the maximum signal power is 1/8 for this IV system, 
 for the maximum signal to noise ratio S/N, 
 
 S/N = | (NF S ) 2 (10) 
 
 A straightforward slope overload criterion is possible for 
 the BSR integrator system because the input must be maximum amplitude 
 limited. The maximum slope of a bandlimited input is dependent upon 
 the maximum slope possible at the output of the bandlimiting filter 
 with some finite amplitude input. In the case of cascaded RC sections, 
 the maximum possible slope occurs with a step input. Specifically, for 
 step size 1/N and normalized sampling frequency F = f_/f r with 2nf 
 being each RC pole, 
 
 1 . e" 27Tf s - 2TTF s e _2TTF s - (2ttF s ) 2 \r e" 2 ^ ... < I (11) 
 
 or, where m is the number of identical cascaded RCs, 
 
 1 - e" 27TF s I (2 F s )'/;: < A (12) 
 
 i=0 IN 
 
 A more meaningful F s for higher order filters is the ratio of 
 
 f s to the -3dB point of the overall filter. For m cascaded RCs with 
 
 an overall half power point of f c ' , 
 
(1 + (f c 7f c ) 2 ) m/2 = 2l/2 (13) 
 
 or 
 
 f c ' = f e ( e ln(2)/m - 1) 1/2 (14) 
 
 Equation 12 can be used substituting F s ' for F s where F s ' = 
 F s x (e n ^ ' /m - 1) ' . Figure 3 shows the curves for m = 1 and 
 m = 2 with F s ' instead of F s . The value of N is the maximum possible 
 where the only source of noise is granular noise. 
 
 Figure 2e is a modification of Figure 2d involving an additional 
 transmission level yielding synchronization and bandwidth advantages. 
 It shall be referred to as Biburst - uncompacted bursts with bidirectional 
 (plus, ground, minus) levels of transmission. 
 
 A Biburst system transmits a bit only when the input differs 
 from the BSR value by more than half a step. The receiver therefore 
 does not need any clock at all unlike the previous systems. It 
 derives its clock from the edge of the input pulses and if nothing comes 
 in, it holds its current value. 
 
 The average bit rate requirement is obviously reduced for 
 Biburst systems. For a IV signal with cutoff frequency f c , the average 
 
 r r 
 
 absolute value slope is f c . With step size 1/N, the system need transmit 
 only N bits per cycle. A normal delta mod system would transmit F s = fs/fc 
 samples per cycle. Therefore, a Biburst system uses only N/F s x 100 percent 
 of the bits of a standard delta mod system under these conditions. For 
 example, with F s = 64 and N = 10 (satisfying Figure 3 first order RC), 
 The Biburst bit rate is 15.6 percent of the maximum. 
 
 This savings figure refers to the average bit rate, not the 
 bandwidth or maximum bit rate of a channel. Multiplexing multiple Biburst 
 systems over a channel to take advantage of this savings is a possibility. 
 
10 
 
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11 
 
 Martin Newman of the Information Engineering Laboratory is investigating 
 multiplexing multiple stations over a channel using amplitude addressing 
 where a specific station transmits pulses of specific amplitudes. 
 
12 
 
 3. BURSTCOMM I 
 
 3.0 Burstcomm I Description 
 
 Burstcomm I (see Figure 4) was a successful attempt to demonstrate 
 the feasibility of using burst techniques to transmit a video signal with 
 readily available integrated circuitry. It also gave a graphic demonstra- 
 tion of digital thresholding noise immunity. 
 
 The video portion of the composite video signal was burst encoded 
 at a 50 M Bit rate using a nonsample/hold 10 bit ramp encoder. The bursts 
 were transmitted along a twisted pair to the decoder--a 10 bit BSR. Noise 
 in the form of periodic pulses was injected into one line of the twisted 
 pair. This system was compared with straight co-ax video transmission 
 with identical noise. The pulse height was adjusted to be equal in 
 amplitude to the height of the video portion of the composite analog signal 
 
 The sync portion of the composite signal was clipped off and 
 transmitted separately from the video portion. This was necessary for a 
 reasonable visual demonstration so that sync was kept through all levels 
 of noise. When sync is lost, picture deterioration comparisons become 
 impossible. 
 
 Not encoding the sync also allowed all 11 levels of quantization 
 
 to be used for grey scale. ECL 10,000 series ICs were used throughout for 
 
 2 
 two reasons: ease of use at 50 M Bits relative to Schottky T L and the 
 
 fact that a wery high speed comparator (10116 line receiver) is a standard 
 
 10,000 series chip. 
 
 A BSR was made with shift registers (10141) with high speed PNP 
 
 transistors (MPS 3640) at each output (Figure 5). Since ECL output 
 
 voltages range from -.9 to -1.8 volts and the buss sum voltage was set 
 
 well below that (-5 to -4 volts), the transistors were operated in the 
 
13 
 
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 active region only for maximum speed. The buss sum voltage was level 
 
 shifted up to ~ 2v DC average to be compatible with a 10116 input. 
 
 The encoding and decoding BSRs were identical except that the 
 
 encoder had 9 bits to the decoders 10. Each fit into a 2" x 1 1/2" x 
 3 1/4" box. 
 
16 
 
 3.1 Burstcomm I Results 
 
 Figures 6 through 9 give an Idea of what quality of transmission 
 was attained and how clearly digital thresholding was demonstrated. Note 
 that the ramp encoder used was without sample/hold and hence was subject 
 to significant slope overload on the non-bandl1m1ted signal. Nevertheless, 
 a usable result was obtained with the superior noise Immunity of digital 
 signals and the simpler circuitry of burst encoding. 
 
 Burstcomm I was demonstrated to ONR in November 1975 1n 
 Washington, D.C. 
 
 Figure 6. Co-Ax Video, No Noise 
 
 Figure 7. Burst Video, No Noise 
 
17 
 
 Figure 8. Co-Ax Video, Severe Noise 
 
 Figure 9. Burst Video, Severe Noise 
 
18 
 
 4. BURSTCOMM II 
 
 4.0 Burstcomm II Description 
 
 Burstcomm II was designed and built as an improved version of 
 Burstcomm I using the same basic circuitry. With the addition of one 
 extra transmission level, sync information as well as an audio and a sensor 
 line are added (Figure 11). The resulting multiplexed signal is trans- 
 mitted along an optical fiber to gain total EMI immunity. 
 
 Specifically, positive going pulses are transmitted for video, 
 audio and sensor information while negative going pulses indicate 
 horizontal and vertical sync as well as the beginning of audio and 
 sensor bursts. Audio bursts are sent every H sync pulse (15.75 KHz) and 
 sensor bursts every V sync pulse (60 Hz). During these times the video 
 signal is blacked out anyway and need not be transmitted. 
 
 Circuit and timing diagrams of the multiplexer and demultiplexer 
 are in the appendix as Figures AD-1 through AD-4, pp. 32-35. 
 
 The multiplexer and demultiplexer are housed in two boxes (see 
 Figure 10) which contain all encoding and decoding circuitry including audio 
 
 filter/amps, video filter/buffers and sync generators. They do not contain 
 power supplies or the system clock generator. 
 
 Figure 10. Burstcomm II 
 
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20 
 
 4. 1 Audio and Sensor Circuits Design Considerations 
 
 Due to the high--50 M Bit--bit rate, the audio and sensor bursts 
 can be treated as sample and hold samples. No slope overload can occur 
 in the 200 ns (10 bits x 20 ns per bit) needed to generate each burst 
 with the bandlimited audio and the nearly DC sensor. Specifically, 
 from Figure 3 for 10 bit bursts, F s > 60 for first order RC filtered 
 inputs. Since the filter used (see AD-5) is greater than first order 
 and F s = 50 MHz/ (10 x 3.2KHz) = 1563, the audio samples are clearly 
 without slope overload. Since the audio is sampled at 15.75 KHz, the 
 Nyquist criterion dictates a cutoff of 7.87 KHz before aliasing occurs. 
 
 To transmit a voice grade audio signal a bandwidth of 3.2 KHz 
 is sufficient. To reduce quantization error it is desireable to encode 
 higher frequencies as well as providing that they can be filtered out 
 and that they will not be folded into the desired frequency range. The 
 cutoff frequency then for the encoder filter is 15.75 - 3.2 = 12.55 KHz. 
 Aliasing which occurs from 3.2 KHz to 12.55 KHz does not matter since 
 those frequencies will be filtered out at the decoder end with a 3.2 KHz 
 low pass. Figure 12 illustrates this. 
 
 Since real filters do not have the abrupt cutoff characteristics 
 of ideal ones and components for filters come in certain convenient 
 values, the actual filters constructed were a 5.-5 KHz LPF in the encoder 
 and a 3.2 KHz LPF in the decoder. Figures AD-5 and AD-6, pp. 36-37, 
 
 circuits. 
 
 The encoder LPF is acutally one-third order constant-k ir 
 
 12 
 section cascaded with two RC sections. The "faithless constant-k" was 
 
 used because of its simplicity and the fact that the filter characteristics 
 
21 
 
 3.2 KHz 
 
 LPF 12.55 KHz 
 
 f $ = 15.75 KHz 
 
 ALIASING 
 
 SAMPLED 15.75 KHz 
 
 
 i 
 
 
 LPF 3.2 KHz 
 
 
 
 VOICE 
 
 1 to 
 
 
 
 f s /2 
 
 
 
 '■ 
 
 12.55 KHz 
 LPF 
 
 
 15.75 KHz 
 SAMPLER 
 
 
 
 3.2 KHz 
 LPF 
 
 ► OUTPUT 
 
 
 
 
 DEAL 
 
 5.5 KHz 
 
 Ih^ 
 
 ACTUAL 
 
 3.2 KHz 
 
 1-, 
 
 INPUT 
 
 15.75 KHz 
 SAMPLER 
 
 3.2 KHz 
 LPF 
 
 OUTPUT 
 
 Figure 12. Audio Filtering 
 
22 
 
 were not critical. With the two RCs , ripple in the passband is less 
 than 1 dB and attenuation at the critical 12.55 KHz frequency is 
 greater than 20 dB. 
 
 The decoder LPF consists of a 3.2 KHz third order constant-k 
 low pass cascaded with a low Q 7.9 KHz trap. The low Q trap flattens 
 out the constant-k response as well as making sure that granular noise 
 at f s /2 = 7.9 KHz is down more than 20 dB. 
 
 The encoder circuit is also a pre-amp which amplifies an 
 input from a high Z microphone to IV levels for the comparator. The 
 decoder includes an audio amp which runs off of 5.2 V and provides a 
 3V output across an 8 ohm speaker (1/4 watt RMS). 
 
 The decoder consists of a counter/D-A converter instead of a 
 BSR to reduce the package count. Due to the fact the audio bursts 
 arrive as separate burst samples there is no need for the "window" 
 characteristic of a BSR. The sensor decoder is also a counter/D-A pair. 
 
 The D-As consist of PNP transistors wired up just like a BSR 
 except that the emitter resistors are weighted according to the binary 
 weight of each counter output. Due to the range of currents (8:1 for 
 a four bit counter) the resistor values had to be adjusted for varying 
 base emitter voltage drops. The final D-A is accurate to within 10 percent 
 of the least significant bit. 
 
23 
 
 4.2 Video Circuits Design Considerations 
 
 The video circuitry (see Figure AD-7, p. 38) consists of filter- 
 buffers at the encoder and decoder. Again the third order constant-k 
 filter is used, this time with a cutoff of 2.4 MHz. This is near the 
 upper frequency possible with effective samples at 5 MHz from the Nyquist 
 criterion. At this point however, there is significant slope overload 
 error of the time/voltage type of the non-sample/hold ramp encoder. 
 Future improvements could include the addition of sample/hold circuitry 
 to eliminate this error. Nevertheless, Burstcomm I showed a useable 
 visual result is possible even with this error. 
 
 The encoder is adjusted to not encode any of the sync portion 
 of the composite video signal since the sync is already being transmitted 
 as negative going pulses. Specifically, two 50 MBit negative going 
 pulses in a row signify HSync and one followed by nothing signifies 
 VSync. 
 
 Sync circuitry (also Figure AD-7) included two Schmidt trigger inputs 
 
 to get clean pulses at ECL logic levels and two monostables at the output. 
 
 As soon as incoming negative pulses are decoded into vertical or horizontal 
 
 sync information, the correct monostable is triggered. 
 
24 
 
 4.3 Optical Link Design Considerations 
 
 To achieve picture quality similar to Burstcomm I, the 
 optical link needs to transmit bits at a rate of 50 Mbits/sec. This 
 dictates a minimum pulse period of 20 ns. In addition, three levels 
 of intensity have to be sent. The following discussion will refer to 
 Figure AD-8 on p. 40 which contains a schematic of the optical link. 
 
 Bifurcated Dolan-Jenner E624 glass fiber bundles are used for 
 a number of reasons; a large diameter (.044") at the transmitting 
 end with consequent efficient coupling in of light, ease of mounting, 
 two LEDs transmitting to one receiver for more total optical power, and 
 most importantly--the fact that a few surplus E624s were floating around 
 the lab. A figure of 45 percent transmittance is given for the 2 1 fiber 
 bundle at around 650 nm. 
 
 Dial co 521-9195 red LEDs were chosen for their relatively 
 high power (for a visible LED), narrow beam (half power into half 
 angle 35o), fast speed of response (1 ns rise and fall time in spec, 
 sheet), and low capacitance (20 pF at V = 0). They are driven with 
 current sources to maximize speed of response. An MPS 3640 PNP is 
 used as each current source at the output of an ECL gate and provides 
 a current swing of 10-40 ma through the LED. Measured optical power 
 output varies from roughly 9 to 31 ywatts. 
 
 The receiving end presented many more design difficulties. 
 An initial design using cascaded cascode stages proved too unstable 
 
 and also lacked sufficient speed. The final simple photodiode-amp 
 
 13 
 combination was derived from a circuit used by a group at GE. The 
 
 amplifier consists of multiple 1/3 10116 line receivers operating between 
 
 ground and -5.2V with 10K resistors for negative feedback stability. 
 
 Only two of the three comparators on the first stage IC are used as the 
 
25 
 
 use of all three cascaded leads to high frequency oscillations. 
 
 The amp operates from DC to around 80 MHz with a measured rise and fall 
 
 time of 6 ns and a gain of 4.7 volts/yamp small signal. 
 
 The photodiode used is an HP4207 PIN photodiode. Advantages 
 of a PIN over an avalance photodiode include decreased sensitivity to 
 power supply noise, increased temperature stability, and lower power 
 
 supply voltages. The major disadvantage is less output current for a 
 
 14 
 given light intensity. Since Burstcomm II was intended as a demonstra- 
 tion with short fiber lengths, the PIN photodiode was chosen. Calculated 
 
 p 
 current responsivity at 640 nm was 3 yamp/mwatt/cm . 
 
 Total optical power output swing of the LED is 22 ywatts. 
 For .1" spacing between emitter and fiber and .022" fiber radius, the 
 solid half angle is 12.50. Using the Dialco spec, sheet, roughly 15 per- 
 cent of the output power goes into the fiber (ignoring coupling losses 
 from reflection). Flux at the output of the fiber (radius .0625") then 
 is 22 x .15/((.0625/2) 2 x it x 2.54 2 ) ywatt/cm 2 = .167 mwatt/cm 2 . The 
 output of the photodiode (again ignoring losses between fiber and 
 photodiode) is 3 x .167 = .5 yamp. The amp output then is 4.7 x .5 = 
 2.35V . Since the output is clipped at ECL logic levels (-9 V DD ) 
 these calculations indicate that the optical link should work. 
 
26 
 
 4.4 Burstcomm II Results 
 
 Figure 13 shows a test pattern transmitted through Burstcomm 
 II. The effects of quantization are clearly visible but the result is 
 definitely useable. Sync was transmitted with no problems. 
 
 Figure 13. Burstcomm II Video 
 Audio presented some problems initially with the sensitive mic 
 preamp picking up radio stations. An RF bypass capacitor on the input 
 eliminated the problem. The amplifier on the demultiplexer causes inter- 
 ference with the video when set to maximum volume. This problem would be 
 eliminated by increased power supply filtering or by separating the 
 amplifier from the demultiplexer. Figure 14 shows what a transmitted 
 1 KHz sine wave (unfiltered) looks like. The top trace is the input, the 
 middle trace is the unfiltered output, and the bottom trace is the difference 
 between the two. 
 
27 
 
 ^•///'-^v 
 
 \v 
 
 Figure 14. Burstcomm II Transmission 1 KHz Sine Wave 
 
 The overall audio frequency response is shown in Figure 15. It 
 
 corresponds well with what was predicted from the design. The actual 
 
 -3 dB points are approximately 400 Hz and 3 KHz. 
 
 -4 
 
 -8 
 
 o -io 
 -12 
 -14 
 -16 
 
 
 
 
 
 
 — •— - 
 
 »s 
 
 
 
 
 
 
 
 
 
 
 \ 
 
 
 
 
 
 
 
 
 
 
 \ 
 
 L 
 
 
 
 
 
 
 
 
 
 
 \ 
 
 
 
 
 
 
 AUDIO FREQUENCY 
 RESPONSE 
 
 \ 
 
 
 
 
 
 
 
 
 
 
 \ 
 
 
 
 
 
 
 
 
 
 
 
 \ 
 
 
 
 
 
 
 
 
 
 
 \ 
 
 
 
 
 
 
 
 
 
 
 \ 
 
 
 
 
 
 
 
 
 
 
 
 
 .2 
 
 .4 .6 .8 1 2 
 
 FREQUENCY IN KHz 
 
 4 5 
 
 Figure 15. Burstcomm II Audio Response 
 
28 
 
 5. BIBURST 
 
 A demonstration of the Biburst encoder (see Figure 2e) was 
 built and tested in the summer of 1975. It was designed to encode/ 
 decode audio frequency signals and transmit them with 3 levels of trans- 
 mission (positive and negative going pulses). 
 
 Circuit diagrams for the encoder are in Figure AD-9. The decoder 
 consists simply of the bidirectional shift register with a clock derived 
 from the delayed edge of the input pulses and mode controlled by polarity 
 of the input pulse. Note that to have no clock on the decoder, the bits 
 must of course be transmitted in return to zero notation. 
 
 Visual results (on an oscilloscope for the encoding of a single 
 frequency) were quite positive for non-filtered output due to the nature 
 of Biburst error. The encoded signal appears only to lag an input by 
 a phase difference dependent on the slope of the input unlike conventional 
 delta-mod which, in regions well away from slope overload, jumps around 
 on both sides of the input. Filtered errors should be similar for the 
 two systems. 
 
 Figures 16 and 17 give an idea of what Biburst encoding 
 looks like for two values of F . Even for low F , degradation of encoder 
 performance is controlled and gradual and a useable result (even though 
 it is attenuated) is achieved. The top trace is the input, the middle 
 trace is the Biburst non-filtered output, and the bottom trace is the 
 difference between the two. 
 
29 
 
 Figure 16. 
 
 F = 100 Noise Amplitude Doubled 
 
 Figure 17. F = 20 Noise Amplitude Normal 
 
30 
 
 6. CONCLUSIONS 
 
 The demonstrations of Burstcomm I, Burstcomm II, and Biburst 
 clearly show the possibility of using burst techniques in the transmission 
 of digitally encoded analog signals. Depending on the specific applica- 
 tion, a version of encoding using burst techniques may prove to be 
 otpimal by taking advantage of circuit simplicity, minimal synchroniza- 
 tion, easy intrinsic companding, or multiplexing possibilities. 
 
 Ramp encoding with sample/hold circuitry on the input is 
 generally the best burst system in terms of bandwidth and could be 
 optimal if the added complexity of the sample/hold circuitry is permissible. 
 Against standard PCM it offers synchronization advantages versus higher 
 bandwidth requirements. 
 
 The absolute simplest system is the original ramp encoder but 
 it requires high bandwidth if its form of slope overload error is to be 
 avoided. If bandwidth is sufficient or the error can be tolerated to 
 some extent, it does provide the cheapest encoding method. 
 
 If companding is useful (due to amplitude distribution characteris- 
 tics of a particular input), a bidirectional delta-mod like system could 
 be optimal. It offers minimum sample bit rates for certain inputs (see 
 Figure 3), those with strongly limited maximum slopes. The major dis- 
 advantage is that bidirectional shift registers versus sample/hold 
 circuitry for the s/h ramp encoder are needed. 
 
 For ultimate lack of synchronization (no decoder clock) and 
 reduced average sample bdt rate, Biburst stands out. Multiplexing 
 possibilities, no sync, and easy companding must be weighed against 
 multiple levels of transmission and increased circuit complexity. 
 
31 
 
 APPENDIX 
 CIRCUIT AND TIMING DIAGRAMS 
 
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42 
 
 LIST OF REFERENCES 
 
 Poppelbaum, W. J., Appendix I to "A Practicability Program in Stochastic 
 Processing," Department of Computer Science, University of Illinois, 
 March, 1974. 
 
 Mohan, P. L. , "The Application of Burst Processing to Digital FM 
 Receivers," Report UIUCDCS-R-76-780, Department of Computer Science, 
 University of Illinois, January, 1976. 
 
 3 
 
 Pleva, R. M. , "A Microprocessor-Controlled Interface for Burst Pro- 
 cessing," Report UIUCDCS-R-76-812, Department of Computer Science, 
 University of Illinois, July, 1976. 
 
 4 
 Taylor, G. L. , "An Analysis of Burst Encoding Methods," Report 
 
 UIUCDCS-R-75-770, Department of Computer Science, University of Illinois, 
 
 December, 1975. 
 
 5 0'Neal, J. B. Jr., "Delta Mod Quantization Noise," The Bell System 
 Technical Journal 45 No. 1 (January 1966), pp. 117-141. 
 
 Protonotarious, E. N. , "Slope Overload Noise in DPCM Systems," The 
 Bell System Technical Journal 46 No. 9 (November 1967), pp. 2119-2161. 
 
 Van De Weg, H. , "Quantization Noise of Delta Modulation," Phillips 
 Research Reports 8 No. 5 (October, 1953), pp. 367-385. 
 
 o 
 
 Bennett, W. R. , "Spectra of Quantized Signals," The Bell System 
 Technical Journal 27 No. 3 (July 1948), pp. 446-472. 
 
 g 
 
 Goodman, D. J., "Delta Modulation," The Bell System Technical Journal 
 48 No. 5 (May, 1969), pp. 1197-1218. 
 
 Taub, H. , Principles of Communication Systems , McGraw-Hill Book Co., 
 New York, 1971. 
 
 Faiman, M. , ed. , "IEL Quarterly Technical Progress Report," Department 
 of Computer Science, University of Illinois, May-June 1975. 
 
 12 
 Lubkin, J. Y. , Filter Systems and Design , Addi son-Wesley, Reading, 
 
 Mass. , 1970. 
 13 Schmid, H. , "Fiber-Optic Data Transmission," Electronics 49 No. 18 
 
 (September 2, 1976), pp. 94-99. 
 
 Compton, R. D. , ed. , "Fiber Op 
 No. 6 (June, 1975), pp. 14-24. 
 
 Compton, R. D. , ed. , "Fiber Optics," Electro-Optical System Design 7 
 
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 Burst techniques are considered as possible alternatives to conventional 
 ones in the realm of digital transmission of analog signals. Three demon- 
 stration systems using various Burst techniques are discussed in detail. 
 
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